I had planned to delete the .com site and just leave the 'free' Angelfire site as a permanent version. Then there were problems accessing Angelfire, and still today I find clicking on any of the links there produces a full-screen pop-up window, which AVG Antivirus warns has some form of malware which it insists on putting in quarantine. I need to find a third, free and permanent solution. For now only the .com site is being updated.
I still get occasional questions regarding circuit boards for the MJR7 amplifier, and a few years ago I did make and sell a few, but only about 30, the design has never been highly popular. That is not unexpected, I intentionally included a number of 'unfashionable' features such as high global feedback, output capacitor and so on. Anyway, my UV light box disintegrated, and since then I just use an etch-resist pen to make any boards I need for new projects. The website is now aimed purely at DIY enthusiasts who can make their own boards. Some have designed their own board layouts, and are happy with the results, but of course all my test results only apply to my own board layout.
WARNING - The lateral mosfets are becoming more difficult to find, and some suppliers are selling fakes. Some examples include cheaper vertical mosfets re-marked as lateral types, and these have a different pin layout with the drain in the middle instead of the source. Both types have an internal rectifier between drain and source which is reverse biased in normal use, but with the fakes these may conduct and at best just blow the fuse. Using a reputable supplier doesn't guarantee no fakes will ever get through, but at least there should be a dependable returns policy.
These are pages which currently only have links from this 'Latest News' page:
Guitar Amplifier With 'Flanger/Echo'
Class-B Feedforward Amplifier.
Boxsim Simulation Speaker Design Example.
A Simple Amplifier Test Method.
Mosfet Amplifier Designs.
Speaker With Conjugate Network.
Transmission Line Speaker Project.
MC Phono Preamp.
There is a useful feature of my mosfet designs, that there is a built in error extraction function. Being an inverting amplifier the feedback network adds the input signal and the inverted output to give almost total cancellation, reducing the undistorted component by 80dB, or even more with a little trimming, leaving noise and distortion in addition to the low level uncancelled music. I included an example for my old MJR6 design, just a 10 sec extract from 'Year 3000' which was the track found to have the highest slew rate when I was checking CD sources for the maximum levels. That sample was amplified considerably (probably 40dB) but still was almost inaudible using my own computer and headphones. I mentioned a 'normalisation' function (18-July) in the Audiograbber program, so I tried increasing the extract to 70% peak level, but it still seemed quiet. Opening it in 'Audacity' revealed a big DC component, so obviously the original sample had a small DC offset. Adding a high-pass filter effect eliminated this, and now here is a new version of the MJR6 error signal:
This will not work if your browser doesn't support the 'audio' tag, instead try clicking this LINK. The noise is clearly audible, but the uncancelled audio is still undistorted enough to easily follow the lyrics, so even amplified to this level any distortion component is still small compared to the noise. Doing the amplification digitally is not ideal, any quantisation distortion will also be amplified, the amplification should have been done with the original analogue signal, but even so the result is fairly conclusive. The noise hopefuly was enough to add adequate 'dither' and avoid quantisation effects, but as mentioned in the earlier link the noise is not the total amplifier noise because the 2uV added by the input resistor is nulled along with the input signal.
Ideally I would want to include some reference level to give some indication of the levels involved, the noise is not very helpful for this because as mentioned it is not the total amplifier noise. As it stands the extract just demonstrates what the signal sounds like if the 'reduced noise' and distortion are both increased by over 10,000 times relative to the original undistorted input. I think nothing useful would be added by repeating the experiment for the later MJR7, the noise will be about the same but the distortion even lower.
I have been looking at the latest book from Douglas Self, Electronics For Vinyl and was pleased to see my version of the vertical rumble filter included as Fig 12.50. As he says the load impedance needs to be high if we want the most extended low frequency response, but using 2u2 capacitors and 5k6 crossfeed resistor I found less than 0.5dB attenuation of the wanted signal at 10Hz with a 20k load, so not a big problem. I have only looked at the free preview at Google Books so far, but it looks good, and can be recommended for anyone not already familiar with the problems of phono preamp design. Those of us who think we already know it all are still likely to find something new, but may be discouraged by the price. To me the greatest surprise is that it is even possible to write 344 pages about 'Electronics for Vinyl', but it does start at a very basic level, and has a lot about the properties of passive components and opamps. It is suggested that polyester capacitors are unsuitable because they produce as much as 0.002% 3rd harmonic distortion with 10 Vrms applied. There is then an interesting observation that the distortion only triples for a doubling of signal level rather than the expected quadrupling. The main reason I didn't use polyesters in my own preamp is that as far as I know they are unobtainable with 1% tolerance.
A question arose concerning my phono preamp. The thing is that the feedback goes to the source of the input jfet, and there is a resistor, e.g. 100R from that point to earth. The suggestion was that the 100R is in effect in parallel with the source impedance of the jfet, which may be about 50R, and so the feedback ratio should include that. The jfet source impedance is given by 1/gm which can vary significantly between different samples of the jfet, so if that had to be included there could be a noticeable gain difference between the two channels.
The signal fed back includes an undistorted component plus a low level distortion component. For that distortion component the low jfet source impedance is as suggested in parallel with the 100R, and the gain round the feedback loop is significantly affected by the gm of the jfet. For the undistorted component however the gate and source signal voltages are almost exactly equal, the difference being inversely proportional to loop gain, and so with high loop gain the source to gate signal is small and very little signal current is driven into the source. Then the jfet source impedance is in effect high, and closed loop gain is determined almost entirely by the feedback network, as we would normally expect. So, no problem. The jfet gm value still has an effect on the open-loop gain, but far less on the closed-loop gain.
The idea that different components of the signal can see different impedances at the same point in the circuit is just a consequence of multiple signal sources. To take a trivial example suppose we have a 1k resistor and apply Vs to one end and Vs + Vx to the other end (both relative to earth), then the voltage across the resistor is just Vx and the current is Vx/1k. For the Vx component the impedance is 1k, but the Vs component is equal at both ends of the 1k and causes zero current, so in effect it sees infinite resistance. In a feedback amplifier we have the input signal and the signal fed back from the output, so there is room for confusion when talking about input stage impedances.
A quick calculation for the phono preamp. With 1V input at 20kHz there will be close to 1V across the 100R gate to earth resistor (I actually split the 100R into 78R plus 22R in the final circuit, but let's keep it simple). The AC current through the 100R is therefore 10mA. Calculating the AC signal current into the jfet source we get around 10uA, about 1000 times less. With 1V at the source and a current of 10uA the equivalent input impedance into the source is 100k, so in parallel with the 100R resistor the effect is insignificant, just a reduction of 0.1% in the resistance of a 1% tolerance component.
Update: This piece about my phono preamp has some tenuous relevance to a lengthy thread on diyAudio about 'current feedback' amplifiers. That's an argument I believe is always worth avoiding, the alleged distinction between VFA and CFA is something I have never found either interesting or useful. The last time I looked there was an argument involving an input buffer with zero output impedance and zero input signal. Yes, under those circumstances any feedback is entirely current. Under different circumstances it isn't. To take an equally unrealistic example, in this case with an input signal and the feedback network connected, then with an infinite loop gain the current into the inverting input is zero, so no current feedback. To stick to a real example, my phono preamp with 1V input at 20kHz has 10mA AC signal in the feedback network and only 10uA into the jfet source, so in that respect not radically different to a conventional VFA. For the phono preamp high loop gain is used for accurate RIAA response, and the single input jfet is chosen for low noise and simplicity.
A useful program I have used a few times is Audiograbber. This has a number of functions, primarily the 'ripping' of audio CDs, so that the tracks can be saved on a computer as wav or mp3 files. One useful additional function is 'normalization' which can adjust the volume of the digital file so that the peak level is some percentage of full-scale. (This is not the same as compression, just a change of volume). The reason why we may want to do this is because of the 'loudness wars' which has led to some recordings going right up to full-scale with no safety margin. This wouldn't matter much except that when converted to analogue in a DAC the result may exceed the 0dB level, an effect known as an 'inter-sample peak'. Some equipment is unable to cope with this and will clip. To put it another way, some badly recorded material played through some badly designed replay equipment may have problems. No surprise there. There are widely varying claims regarding the importance of this effect, but it is easy to check whether there is an audible effect by using the Audiograbber normalization function to reduce the peak level in the digital file. A setting of 90% is probably adequate in practice, taken too far the signal to noise ratio will suffer, but at 90% there is a less than 1dB loss in signal level. There are alternative programs with the normalize function, including foobar2000 and Exact Audio Copy, and some say they are better than Audiograbber, but I never tried these. I just tried comparing a wav file first normalized to 100% then to 90% and then 70%, but it really isn't that easy, unless the track has lots of 100% peaks to start with there may be too few peaks to be a noticeable problem. Also, my current soundcard may have a perfectly adequate overload margin.
Profusion can now supply the Exicon lateral mosfets in selected bands based on the value of Id at Vgs = 0.5V. The lowest (red) band is from 105 to 125mA and the highest (white) from 185 to 205mA. They will not guarantee to supply any requested band, but for a quantity of the same type all will be the same range.
For any of these bands the MJR7 quiescent current can be set to 100mA even with the LED limiting the gate to gate voltage, so this is then certain to not be a problem, and the extra cost of the selected ranges may be worth paying. The unselected range probably includes those rejected as outside the selected ranges, so the chances of getting a high Vgs mosfet may now be higher.
There are still a few pages described as 'work in progress', but that is mostly untrue, I just lost interest or had nothing new to add. These and a few others are incomplete and I hope to update them eventually. One such page is about signal generator design. My conclusion at the point where I stopped was that with sufficiently fine 'tweaking' distortion can be reduced to extremely low levels. Getting easily repeatable low levels from a variable frequency generator is a different matter. One point I intended to investigate but didn't is the optimum jfet characteristics for use as a voltage controlled resistance in the level control. I got as far as mentioning that my choice of the 2N5457 was less ideal than the 2N4091 used by Bob Cordell in a similar but lower distortion circuit. There is probably more to it than just the type, we also have the option of whether to use it at low resistance with close to zero Vgs or at higher impedance with a more negative gate voltage (assuming n-channel). I strongly suspect I got both type and operating resistance wrong, or at least far from optimum. An article from Siliconix, FETs As Voltage-Controlled Resistors says a high Vgs(off) is an advantage for low distortion, so the 2N5457 specified as 0.5 to 6V could be a bad choice.
Updates: There is a very long thread about signal generator design at diyAudio. I don't want to read through the whole thing, but in post 90 Bob Cordell mentions the requirements for jfet selection, and he says high Vgs(off), but also low Rds(on), which may suggest low operating resistance is better. The Siliconix reference, in Table 1, appears to find no advantage in low resistance operation.
In the same thread I also found this link to an article about Amplitude Stability of RC Oscillators which demonstrates that excessively linear amplifiers used in lamp stabilised oscillators can have problems, which reminded me of the unexplained distortion at 100Hz intervals using an LM4562 op-amp in part 3 of my design articles. I'm not entirely convinced, but I really don't care enough to try working through that analysis.
Update 2: I found a similar result described starting here and an explanation in post 6928 that the 100Hz interval components were caused by a nearby DECT base station (cordless phone) which I also have operating within a few feet of where I tested the oscillator. The 55MHz GBW of the LM 4562 may be why this was the only op-amp tested to have this problem.
Also, another recommended jfet is the MMBF4391, or PN4391. These appear to be similar to the more easily available J111.
A Fairchild application note AN-6603 includes distortion plots for various examples. Figs 16-21 show signal levels for fixed levels of distortion as a function of Vgs. Fig.16 for example shows that at 0.5% distortion a 40mV signal level can be used at Vgs = 0 but there is the same distortion at only 6mV at Vgs = -2.5V, so this shows that low Vgs operation is better for low distortion.
I have added a page with my latest guitar amplifier project. I had doubts about whether to include this, it is really the wrong way to make a 'flanger effect', but it is perhaps an interesting example of a PT2399 time delay circuit:
Guitar Preamp and 'Flanger' Using PT2399 Time Delay IC
My old freeserve email address will no longer work from may-2017, the service has had a series of owners, starting as part of a dial-up service run by Dixons and ending up being discontinued by EE. It was never very reliable, I often had problems to log in. Now I am setting up a gmail address.
Apart from that nothing much is new, I have just finished making a guitar amplifier starting with an 'ADS mixer amplifier' I bought on eBay. I removed everything apart from the transformer and bridge rectifier and added a fairly basic lateral mosfet amplifier. There is a row of 7 controls on the front panel, and I want to use 2 for a 'flanger' type effect. Many years ago I experimented with a reel to reel tape recorder using two replay heads connected in series and with a mechanical lever to vary the length of tape between the heads. The result was impressive, but looking at current designs it is invariably stated that the time delay between the two added signals should be just a few msec for this effect. There is an easily available and low cost time delay IC, the PT2399, which looks interesting, but has a minimum delay around 30ms, which may be a problem. My tape effect certainly had a far higher delay and still produced a good result, so some experimentation seems worthwhile. A good source of information is Project 26A on the Elliot Sound Products website, which also says it is unsuitable for use as a flanger. There are a few examples of flangers using this device, but they use a pair for the two signal paths so that the delay difference can be small or pass through zero.
I promised to think about add-on circuits to make the phono preamp suitable for moving-coil pickups. I don't have a MC pickup myself and don't plan to build anything, but I have added a page, MC Phono Preamp with a few not very original ideas on this topic.
I found a page on the TT Electronics Semelab website from Aug 2016 announcing the discontinuation of 'Magnatec' products. The list linked to includes the ALF and BUZ series lateral mosfets and also the 2SC2240BL I still recommend for my MJR7 amplifier. The Farnell UK site also lists the ALF mosfets as 'no longer manufactured'. There are of course alternatives still available, but maybe not for long. I still don't want to resort to vertical mosfets, they add a few problems, but the time may come when that is the only option.
Best wishes to everyone for a happy and peaceful new year.
In my phono preamp design pages I mentioned an optional time-constant of 3.18usec, often referred to as the 'Neumann pole' on the grounds that it is sometimes claimed to be included in the pre-emphasis circuits of the Neumann disc cutting lathes. This is widely disputed, and for example here in Neumann's Last Pole is a circuit diagram of the Neumann SE66 recording equaliser, which uses a second-order low-pass filter at 33kHz. The later model SAB-74B is said to have a similar filter at 50kHz. This will have little effect at 20kHz, certainly less than the disputed first-order 3.18usec filter would have, so including this time-constant in the RIAA correction will not be an accurate correction. I mentioned my own preference to leave out the 4th time-constant, and this is now a positive recommendation. In the discrete component circuit the 75pF should be included in the output section, and in the single op-amp circuit the added low-pass filter at the output is included. The simple non-inverting pre-amp has a 4th time-constant whether we want it or not, so the 3.18us value seems to be as good a choice as any, and needs to be calculated anyway so that it can be cancelled by an additional first-order low-pass filter, so we still can't avoid the slightly tricky four time-constant calculation. Fortunately that calculation has been done and published by a few writers, including myself, so provided the 3.18usec 'standard' is adopted there is no need to recalculate everything for every new design.
I found another error in my phono pre-amp layout diagrams, one of the electrolytics was reversed, and now I have corrected this, and marked the component with a red star. The capacitor would certainly be damaged by being reverse biased, so if anyone has used this layout that capacitor needs to be replaced. A worse problem would be if anyone assumed the capacitor to be correct and connected the power supply to bias it correctly, then serious damage to many components would be likely. The correct power supply voltages are shown in the final diagram.
I have also added a circuit diagram showing the voltages at all points in the circuit, which may be helpful for fault-finding.
I downloaded Visaton's free Boxsim program, said to give accurate simulation results including baffle effects. I already have a pair of Visaton FRS8 full-range speakers which could be used with the W170S, maybe crossing over around 400Hz.
I have worked out a few examples using the FRS8 and W170S just to learn how to use the program. The predicted frequency responses are surprisingly good, mostly within +/- 1dB apart from at lower frequencies where we probably don't want a flat response, there being some effect from placement near a wall and floor giving bass boost. That is one part I am not yet entirely sure about, the simulation seems to assume a 6dB loss from low frequency diffraction, but examples from the Visaton website seem to aim for a flatter bass response, maybe assuming intended use well away from walls. Anyway, here is the Boxsim example. I started with a simple passive crossover design, and also included an active crossover version.
I have added my simulation results for speaker current peaks when driven by a voltage pulse, Speaker With Conjugate Network, showing the results both with and without a parallel conjugate network, and the effectiveness of this addition is demonstrated. One slight surprise is that just adding a RC Zobel to prevent the rise in impedance at high frequencies can actually make the peak current requirement higher. Full level rectangular pulses appear not to be a common feature of recorded music, so in practice there is little to be gained by adding a conjugate network, and the inconvenient component values needed are also discouraging.
Earlier on this page I mentioned a piece I wrote about the Peak Current Requirement of Speakers. This was based on reports in recent audio design books, but I had never seen the original paper refered to, Peak Current Requirement of Commercial Loudspeaker Systems.
Now today I have been given a copy, and have been trying to decipher it. There are a few problems, for example describing the test signals as 'bandwidth-limited' could mean anything from a first-order bandpass filter to a 'brickwall' filter, and the input signals shown in the final results look like perfect rectangular waves. Also it is claimed that most commercial amplifiers are designed and tested for only the specified resistive load, which I doubt was ever true, even in 1987.
Anyway, the clever part is the construction of a test signal for a given speaker designed to give the maximum peak current, which represents the 'worst case' signal. These rectangular wave shapes are however just an extreme example of what we will find with any square or rectangular wave input, for a very simple reason. I repeat what I wrote earlier:
'Starting with a square wave voltage applied to a speaker, this consists of a fundamental frequency and a theoretically infinite series of odd harmonics. The resulting current has the same series of harmonics but because the speaker impedance has reactive components there is a frequency dependent phase shift of these harmonics, which changes both the shape of the current wave and its peak level. The square wave harmonics have the relative phases necessary to give the minimum possible peak amplitude, so almost any change from this gives a higher peak level, so the current wave will invariably have higher peak levels than if the speaker was a pure resistance which would have left the phases unchanged. There is no theoretical upper limit, shifting all phases by 90deg would give an infinite peak amplitude, limited in reality by the finite bandwidth and speakers having less than 90deg phase shift.... The effect applies for any rectangular test voltage, in all cases the phases of the frequency components start with the values needed to minimise peak level, so almost any phase shift applied to the current increases its peak level.'
The most serious objection to this sort of theoretical analysis is that it is so much easier just to play some music through the speakers and monitor the peak current requirement. I gave a reference in my previous piece to an example published in Stereophile, which failed to find any unexpected current peaks, and my own tests also found nothing to be concerned about.
Even if there really was a problem we can avoid it without needing to make an amplifier capable of providing 100A peaks. A far easier solution is to design a network to connect in parallel with the speaker to flatten the impedance curve and so reduce the phase shift. This is called a 'conjugate network', and is often encountered in discussions about current-drive speakers. The current peaks into the speaker would be exactly the same with the troublesome rectangular voltage waves, but most of that current will be provided via the parallel network, not from the amplifier. This is perhaps counter-intuitive, adding an impedance in parallel with the speaker we may expect it to take even more current, but think of a tuned circuit with a parallel inductor and capacitor, at resonance the impedance can be very high, so a driving amplifier needs to provide very little current, but the individual currents through the L and C can be far higher, but in almost opposite phase so that they almost cancel. A thread on diyAudio includes on the first page an example of adding a conjugate network. For complex speakers the network may be difficult to work out, but for the present purpose, with voltage-drive, we don't need to be so exact about flattening the impedance, small errors will have little if any effect on the frequency response. The simplest example is a single drive unit, and here is an example of a calculated network.
UPDATE: I just tried a simulation for a single pulse and a single drive unit, using a very approximate conjugate network, and found that after the end of the pulse the speaker current peaks at -3A, and at the same time the network current peaks at +1.3A, so there is partial cancellation and the total amplifier current only reaches -1.7A. As expected the amplifier current has almost the same shape as the original voltage pulse, so the total load is behaving more or less as a resistance.
The phono-preamp continues to be the most visited page, which is a little worrying given an almost complete lack of feedback from constructors, with my own construction the only fully tested working example I am aware of. I do know of two constructors who had problems. I wanted to add an alternative simpler circuit which I would be happier to recommend, using a dual opamp and single equalisation networks. Why bother with yet another conventional opamp circuit? Well, the vertical rumble cancelling circuit can still easily be included provided the output load is not much less than 10k. Also, I avoided the calculation for a 4 time-constant feedback network in my previous coverage, but it turned out to not be too difficult. The calculation assumed infinite op-amp gain, but compensating for finite op-amp gain is demonstrated in simulations using small adjustments of the component values. I have added a page Op-amp based Phono Pre-amp..
UPDATE: I have worked out slightly better component values and improved simulation accuracy, and the final circuit is now updated to give a theoretical +/-0.01dB accuracy, but the 'tweaks' involve changes comparable to the 1% component tolerances, so the real accuracy is unlikely to be so good, but still more than good enough.
I mentioned a possible transmission line speaker project, and so far I have been collecting information and ideas, and doing a few experiments to compare absorbent materials. The results are not particularly conclusive, but there is no more I really want to do for now. A practical design may emerge eventually, but for now here is an initial page, Transmission Line Speaker Project.
I suggested earlier on this page that two or more of my first-order rumble filter circuits could be used in series to give higher attenuation at 10Hz. This seemed obvious enough not to need a simulation, but today I tried a simulation of another higher order filter, the Oldfield circuit linked to in my references, Stereo Rumble Filter, and found it worked well, but has a rather unfortunate channel reversal effect over a range of frequencies, with an input to only one channel the output is higher from the other channel. This prompted me to do a simulation of two of my first-order circuits in series, and unfortunately this has a similar problem. I have added an update at the end of the Rumble Filter section on the 'Phono Pre-amp Design Theory' page. The Oldfield circuit has the advantage that the resistor values in the filter can be adjusted to minimise the problem, but working out the optimum values may not be entirely trivial, trial and error may be quicker. The problem appears to be further down the frequency range for my two first-order circuits, under 30Hz, so I would not expect a serious audible effect. Few recordings will have significant levels under 30Hz and any such components will probably be recorded in mono, not on just one channel.
Initial simulations suggest that adjusting resistors to reduce the channel reversal effect in the Oldfield circuit also reduces the 10Hz attenuation, so it may be little or no improvement on my in-series first-order filters. Another idea I checked was that if two of my circuits in series produce this reversal effect maybe more stages will reverse it back again, but it doesn't work that way, adding more stages increases the levels of peaks and dips and also moves them higher up the frequency scale, so using more than two stages is not recommended.
The quiescent current preset adjustment in a power amplifier should always be designed so that if the preset wiper becomes open-circuit the output stage current will fall, not rise. Looking at my MJR7 circuit it is not entirely obvious that it will work like that, the mosfet gate currents are very low, so the only thing discharging the 100n capacitor is the leakage current through the zener gate protection. I think that should be enough to guarantee a fall in quiescent current, but I don't have a figure for drain to gate leakage, I assume it is very small, but to avoid needing to worry about that sort of thing we could add something like 5k6 in parallel with the 100n capacitor. I don't find that worrying enough to take my own amplifier apart to add the resistors, but anyone building the amplifier may want to include this modification.
Incidentally, some mosfet output designs use different value series gate resistors to compensate for higher capacitances in the p-channel device. I used 300R for both, but did try different values when measuring distortion without finding much effect. The 300R limits driver stage dissipation under fault conditions when the zeners conduct, so should not be reduced too far.
There is a new article by Douglas Self, The Devinyliser in Vol.11 of Linear Audio. This is an excellent and thorough treatment of the rumble filter technique where low frequency opposite phase rumble signals are cancelled. My own simple version is included in the article together with higher order circuits giving better cancellation at 10Hz. As I mentioned below on this page (17 Jan 2016) things get tricky if you want more than a 6dB/oct attenuation of out of phase bass. The solution in the Self article is to use an all-pass filter in the subtraction circuit, which is highly effective. One point he may have missed, which I also missed myself until today, is that it is possible to use multiple circuits in series to get higher attenuation, but to match his 2nd order result in Fig.12 would need 4 of my circuits in series to achieve -40dB at 10Hz and -12dB at 32Hz, which needs more capacitors and more op-amps, but on the plus side component tolerances may be less of a problem. (But see update above on 07-May-2016).
(Why, you may ask, would it take four 1st order filters to match a single 2nd order? Partly because Fig.12 is actually 3rd order, which is what you can get if you subtract a 2nd order Butterworth from a 1st order all-pass, and partly because of the more gradual initial slope of the 1st order types.)
Anyway, the article is very good, but unfortunately the Linear Audio page says 'download not available', so it seems to be necessary to buy the entire volume to read it. Someone kindly showed me their copy, so I didn't need to.
I mentioned earlier on this page, 21-Aug-2014, US Patent 8466744 which lists my MJR7 amplifier under 'Other references'. There is now a discussion about this patent on diyAudio which reveals some additional information. A commercial design from Bryston is said to be based on the patent, and a representative of that company says the 'novel methods of maintaining stability' are an important feature. The diagrams are described as 'simplified and representative' so there would be no point trying to build or even simulate the circuit. Anyway, if there is anything genuinely new or useful I don't see it.
I remember a quote, probably from Feynman, concerning the laws of electromagnetism, to the effect that 'there are no known non-trivial solutions to these equations'. The static field of a uniform charged sphere is an example of a trivial solution, and this sort of thing is quite common in physics, hence the well known jokes about 'spherical chickens in a vacuum'. The same thing is certainly true of acoustics, and is why trying to design speakers starting from the laws of sound propagation will only be an approximation. There are apparently some good simulations around which give reliable outcomes, but my own preference is to start with a design allowing adjustment of the important variables so that they can be optimised by variation and measurement. The example I included previously was a distributed port bass reflex, and by blocking some of the ports it was easy to find the best balance between distortion and bass response. I mentioned that I wanted to try some sort of transmission line enclosure, and I have a few ideas how to make adjustments without needing to completely rebuild a conventional enclosure, so that may become my summer project.
I wrote previously that to increase the Q of a speaker we can reduce the box size, which is true, but not a very good idea. I tried the simulation at mh-audio using the Monacor SPH135AD specifications, fs=39Hz, Vas=26l, Qts=0.37. What this shows is that with a 10 litre box the output at 40Hz is down at -11dB, and if we reduce the box size to 5 litres the 40Hz output drops almost to -15dB. An alternative approach is to use a speaker with higher Qts. Keeping a 10 litre box and increasing Qts to 0.8 gives 40Hz at -8.7dB and a boost of 4dB at 80Hz (which we would then correct e.g. using a Linkwitz Transform). If we then increased the box to 20 litres then 40Hz is at -4.6dB. Increasing Qts however can have its own drawbacks, it can be achieved by using a smaller magnet, but that reduces sensitivity, so although output at 40Hz may be better relative to output at 1kHz, if 1kHz output is reduced by using a smaller magnet then actual 40Hz output may not be as good as we hoped. The mh-audio simulation only gives relative levels at different frequencies, not actual sound pressure levels, so some further thought is needed to find the optimum design for high bass sensitivity.
High bass sensitivity allows higher level bass before dissipation becomes a problem if we boost the amplifier gain to achieve whatever bass response we want, but ultimately limited by maximum linear cone displacement Xmax. For the SPH135AD Xmax is specified as +/- 1.75mm, which is not great, but maybe it is good enough if we don't need high sound levels. These speakers have been suggested as substitutes for the KEF B110 in LS3/5A copies, but the B110 has Xmax 6mm. The lower Xmax does however allow an improved sensitivity, specified as 89dB/W.
The 'problems' mentioned previously for the speaker bass boost capacitor can easily be avoided using my MJR7 amplifier. The existing output capacitor of the amplifier could be reduced to something like 1000uF so that it cancels some of the series inductance of the speaker just below resonance. At first this looks pointless because the feedback taken from after the capacitor flattens the frequency response. What it does achieve is to reduce the amplifier output voltage prior to the capacitor in that low frequency range, and so gives a greater overload margin so that we could then boost the low frequency gain using a conventional Linkwitz Transform circuit ahead of the amplifier, but with less danger of amplifier clipping. Designing the speaker for a relatively high Q (e.g. 1.2) could compensate for the attenuation at and above resonance, and the Linkwitz Transform can be designed to match that Q value and give whatever final effective Q we require, typically 0.5 to 0.7. Initial simulations suggest the resulting improvement in overload margin could be more than 3dB, equivalent to doubling amplifier power, but only over a small frequency range. Using this approach we don't need that extra capacitor, and the existing one is lower value and cheaper, and also still included inside the feedback loop to reduce any distortion, and also we can still specify a high damping factor. Anyway, it's an idea to play around with, but probably not really useful, few recordings have very high level low bass, so overload is unlikely to be the most serious problem.
I have been reading more about speaker design, and one idea I was reminded of is to add a series capacitor to the bass speaker, typically 1000uF, to cancel some of the series inductance at frequencies just below resonance. This increases the drive voltage at these frequencies, equivalent in effect to boosting low frequency amplifier gain, but able to produce a higher voltage across the speaker before amplifier clipping. Output at and above resonance is however reduced, but this can be compensated to some extent by designing the speaker for a higher Q, e.g. by using a smaller box. I need to do a few simulations to see if this can be made to work well, I have some doubts, but there is an obvious advantage that the speaker is then protected against damage under fault conditions causing high DC levels. There are probably a few disadvantages, not least the fact that some people will worry about passing the signal through yet another capacitor, and others may worry about their damping factor.
An important point is that some speaker designs do use this technique, and they may have some problems when used with my MJR7 amplifier. I mentioned in my setup and testing page that adjusting the output stage operating voltage has a problem with long time constants if there is not a low resistive load, and it would be a good idea to add a parallel resistor, e.g. 47R 5W, if one of these capacitively coupled speakers is used.
My original notes on crossover filters are 15 pages of rather tedious analysis, so I have extracted a few diagrams and added new descriptions to present the essential information in a much shorter and more easy to follow version, with the title Asymmetric Crossover Filters. This is by no means a complete coverage of the many possible variations or uses of asymmetric crossovers, one example I found recently is that it is possible to use asymmetric filters to compensate to some extent for different acoustic path lengths when bass and treble speakers are mounted on the same flat baffle. An interesting account of this can be found at the Parts Express Forum.
Regarding linear phase filter design using time delays, the invention of this technique is sometimes credited to Lipshitz and Vanderkooy, who described it in their 1983 JAES paper, but they don't make that claim, and their own references include two previous publications describing the time delay method, in 1979 by Tanaka and Iwahara and in 1973 by R.M.Golden. There was also a paper by Ng and Rothenberg, 'A matched delay approach to subtractive linear phase high-pass filtering', published 1982. My own work covered only a small subset of these filters, and I had never read any of the others, so I was pleased to be given a copy of the Lipshitz and Vanderkooy paper today, and I have been trying to decipher their mathematical approach. I see how they did it, matching the time delay element to the low frequency group delay of the low-pass stage. My own method described previously leads directly to the filter responses giving a higher order high-pass. These are covered in the L-V paper but their method of analysis is more general and, to me at least, more difficult, but my results agree with their general formula. It is said that the 5th or higher order low-pass results are 'unstable', which I have no reason to doubt. The 4th order low-pass and resulting 5th order high-pass will work, as I suggested, but the responses are plotted in their fig 7b, and clearly have significant peaks in both high and low-pass, so that saved me from wasting time plotting them myself.
I just noticed a discussion on diyAudio (starting around post 354) about the vinyl rumble filter idea where low frequency out of phase signals are cancelled. My own simple first order Rumble Filter (Part 2) is mentioned later and a few other versions. I listed a few examples at the end of my page, but I never bothered to check these in any detail, assuming that if they were published then they must work well enough. Some have now been investigated by Douglas Self, and it appears the higher order types can have problems, including uneven response and sensitivity to component tolerances. Evidently things get tricky if you want more than a 6dB/oct attenuation of out of phase bass, and fortunately I avoided these difficulties. My circuit with the component values shown only reduces 10Hz vertical rumble by about 10dB, but that is still a noticeable improvement.
Update: post 557 in the diyAudio thread shows one of the problems with higher order filters if you want not just a flat gain but also zero phase error, the unwanted signal can only be reduced at 6dB/oct and even worse has a 2dB peak. This has a known solution, but that involves time delay functions. According to Douglas Self ('The Design of Active Crossovers', section 6.2 page 134-140) using a time delay can give a high-pass with the same slope as the low-pass response. I did some work on that problem long ago when I was a student and my unreliable memory is that there exists one low-pass response of any given order which can be combined with a time delay to give a high-pass of one order higher. I must search for my old notes and check that.
Update 2: Yes, my memory is almost correct, there are low-pass responses for which the corresponding time delayed high-pass is one order higher. I worked that out myself for 2nd and 3rd order low-pass. I have done a Google search and find this is well known, but some say this doesn't work for higher orders (not entirely true, see later). The calculation is easy enough, just start with a first order all-pass (a-s)/(a+s) and subtract a low-pass e.g. 1/(1+bs+s2) and the result can then be made into a third order high-pass by choosing a and b to make all the terms in the numerator zero except the -s3, which makes both a and b equal the square-root of 2. Generally starting with a nth order low-pass we get n equations for n unknowns, which at least for n= 2 or 3 is solvable. If we get a higher order high-pass using the all-pass function then that will also be true for a time delay chosen such that the all-pass and time delay become identical as frequency is reduced towards zero, which is also where the ultimate slope of the high-pass is reached. Maybe I will add a scanned copy of my old work, but it probably adds nothing useful to what has been published elsewere.
Update 3: I just tried the calculation for a 4th order low-pass and it seems to work fine, 4 equations for 4 variables and easily solved to give a 5th order high-pass. I've just found more of my old notes, and I already did the 4th order calculation then, with the same result. Only the final slope at low frequencies is predicted, actually plotting the complete responses may reveal problems, I only plotted the 2nd order low-pass and corresponding 3rd order high-pass, and using a time delay that had a 0.5dB peak in the high-pass gain about 2 octaves above the crossover frequency. Using a first-order all-pass instead of a time delay avoided the peak.
On my Links page, in the physics section, I recommended the Susskind Lectures available on YouTube, and having nothing better to do I was working my way through some of these. There is one serious problem I found in the Quantum Entanglements Part 1 series (there is no part 2 and what is listed as part 3 appears to be on a different subject, but that isn't the big problem). Having got through the first 4 lectures, about 8 hours, covering some fairly basic theory we arrive at lecture 5 covering Bell's theorem and other more interesting topics, only to find the video quality to have sunk to the level where little of what is written on the board is legible. It isn't just me, the comments on that page make the same complaint. With some difficulty it is possible to reconstruct the missing graphics from what is said, and I would hate to give up after all that, so I will try to continue to the end. Anyway it is probably not such a good recommendation as I thought.
One easy to follow section concerns Bell's Inequality, showing that it is purely classical and extremely trivial, it can be written in one line and the proof is another single line plus a simple diagram with three overlapping circles. The big deal is that it can be violated by quantum mechanics, and this can lead to some interesting conclusions (e.g. the need to abandon 'local realism').
One feature of my amplifier designs I forgot to cover in the design notes is the phase shift. It was suggested in Wireless World August 1973 (An aproach to audio amplifier design) that power amplifiers should have phase shift at 20kHz no more than 6 degrees, including all previous preamp and filter stages, so the power amp alone should be restricted to about 2 degrees, necessitating a bandwidth in excess of 500kHz. The phase shift of the MJR7 reaches 24 degrees at 20kHz, so why is this not a problem?
There is a rather good article on the Audioholics website, Phase Distortion Audibility, which concludes that under typical listening conditions including room reverberation 'phase distortion' is almost entirely inaudible, but under some other conditions a difference can be reliably heard.
Looking at some of the references there is one important point often missed. Suppose we run a listening test with two sinewaves, 1kHz and 10kHz, and shift the relative phase of the 10kHz by 10 degrees, and find under certain conditions, e.g. in an anechoic chamber, that we can hear a difference. The question is whether this is what an amplifier with 10 degrees phase shift at 10kHz actually does. The answer is almost always no. If we look at the two test signals on an oscilloscope then the wave shape will be noticeably different with the different phase shifts. To return the shape to the original without phase shift there are two ways to do this. We could just remove the phase shift at 10kHz, but we could alternatively add phase shift of 1 degree to the 1kHz component, which again restores the wave shape. With 1 degree at 1kHz and 10 degrees at 10kHz we have a 'linear phase shift', i.e. the phase shift is proportional to frequency, and this is equivalent to a constant time delay, in this case 2.78 usec. That has just the same effect as pressing the 'play' button 2.78usec later, or moving your listening position a little under 1mm further from the speaker, and there is no change to the wave shape, and nothing that could be heard. That is more typical of power amplifier phase response, and looking at the simulated phase as a function of frequency for my MJR7 up to 20kHz it is almost perfectly linear, equivalent to a time delay of 3.3usec.
Of course if we go further up the frequency scale the phase will eventually deviate significantly from linearity, and if we used a test signal with frequency components way beyond 20kHz we could observe some obvious effect on the shape. Hopefully we normally listen to band-limited audio signals, and we don't care too much what happens to the phase shift at 100kHz. I wrote on another old and now deleted page that it is unfortunate that there is such widespread use of square wave testing, this can lead to worries about all sorts of 'problems' such as 'ringing' and 'settling-time'. For example I pointed out that ringing is often interpreted as something undesirable being added to the signal when in some cases there is something undesirable (harmonics above the audio frequency range) removed from the signal. Aiming for a 'nice looking' square wave response may be a recipe for changing the wave shapes of band-limited signals, which is what we really need to care about.
To summarise, the one thing we possibly need to avoid is 'deviation from phase linearity' in the audio frequency range. That, together with a flat frequency response, is what detemines the accuracy of audio wave shape reproduction.
I mentioned a new idea for improving my distortion extraction circuit. The problem with using the signal nulling method is that to test a non-inverting amplifier we need to invert either the test signal or the amplifier output so that we can subtract them to leave just the added distortion. The inverting stage needs to have distortion well below that of any amplifier we may want to test, and that is difficult. Using a low distortion op-amp is no help because we may want to test such an op-amp, so we want our test circuit to have even lower distortion. One idea I want to try is using feedforward error cancellation in the inverting stage, and I have added a page with a preliminary example, Distortion Extraction.
I mostly used this sort of circuit for measuring low level harmonic distortion, but one useful feature is that tests can be done using music signals. Such tests were done many years ago, first by Quad, then by others, and they found nothing audible when listening to the extracted distortion from well designed amplifiers. That was my own conclusion even for the worst of my recent designs, the original MJR6, but it is difficult doing this sort of test when driving a speaker because of the effect of the frequency dependent impedance, and I had to use a more devious method to get good results in that case. The page where I explained how that was done was deleted some time ago, but I found an old backup copy, and with a few updates is now available as A Simple Amplifier Test Method. The final circuit on that page is the one I used to extract the original MJR6 distortion with music driving a speaker. This method only works for inverting amplifiers with a close to unity-gain output available from the input stage. The MJR6 and MJR7 are this type, but most amplifiers are not, so it is not a widely applicable method.
Here is a ten second example of 'distortion' extracted from the old MJR6 as a wav file (1.69MB). Although amplified considerably this is still at a very low level, and it sounds mostly like noise plus a low level of uncancelled music signal. The track used was the one I found when testing CDs for maximum slew rate, this track had the single highest peak I found after a few hours searching, equivalent to a full level 10kHz sinewave. I had a link somewhere to a Stereophile test of the wider bandwidth DVD-A recordings which only found the equivalent of full level at 12kHz.
Although it is reassuring to find nothing worth worrying about in this sort of test, in practice the standard 19+20kHz intermodulation test is more demanding, far easier to perform, and the results more useful for comparing alternative designs.
Apologies for lapsing into theoretical problems, the website statistics reveal that the vast majority of visitors are mostly interested in the practical projects, but I see no point just adding new projects unless there is some vaguely original idea involved, or something I need for my own use.
What is perhaps notably absent is a high power amplifier example. I have never thought high power levels to be a major design challenge, using 2 or 3 parallel output devices usually reduces open-loop distortion and also allows higher feedback because of the lower output impedance and so reduced effect of reactive loads. Anyway, I never tried my MJR11 circuit, and it could be pushed up to maybe 200W using dual die mosfets, so I may at least try a single channel version sometime. I also need to build a new distortion extraction circuit to measure lower levels, and I have an idea how to do that. These things can take a long time, mostly because of inefficiency.
I have finally given up trying to write a page about the output impedance of common-emitter stages, and how it is affected by signal source impedance. The latest version leads to a simple model (actually just a variation of the hybrid-pi model) giving about the right answers, but still not entirely convincing. I have included the page here for anyone interested, but have to advise against wasting much time trying to follow it, it is all rather confused and confusing. It is worth repeating what I wrote earlier on this topic:
If we just use Zo = 1/hoe we may look at the data sheet for hoe and for the BC560B find its typical value to be 10uS, but we also find a maximum value 60uS. The ratio of maximum to typical is 6, far greater than the ratio of maximum to typical current gain hfe, which is just 1.5. The important thing to learn from this is that it is a bad idea to design circuits where either hoe or hfe are critical, which would require testing and selecting transistors. If we want very high output impedance there are more predictable ways to do that, such as adding a cascode (common base) stage.
The output impedance of a common-base stage is also a function of the source impedance from emitter to earth, but that is relatively trivial to calculate, and a high source impedance gives a high output impedance, though it took me a while to understand why with an infinite source impedance the output impedance is only half of Rcb in the hybrid-pi model. (Hint: the collector and emitter currents are not exactly equal).
Regarding the h-parameters, if we accept the ratio of output impedances for source impedance varied from zero to infinity to be 2 at Ic = 1mA, as suggested by my measurements, then only hfe and hre are independent variables, and given their values we can calculate hie and hoe. If so this would confirm that the 'typical' h-parameters found in data sheets are often not a consistent set, as suggested by one of the references I mentioned. We could also start with hie and hoe and calculate hfe and hre. I mentioned a formula for Zo which gave a negative output resistance for the 'typical' h-parameters of the BC560B, but replacing these with a calculated set gives a reasonable positive Zo, so I guess it is the h-parameter sets which are unreliable rather than the equations or the models. Specified h-parameters should evidently be used with caution.
On this page, 08-Jan-2014, I mentioned a US patent for a 'new' feedforward amplifier, which I suggested was nothing more than the Quad current dumping circuit from 1971, originally used in the Quad 405. I now see that an amplifier is available claimed to be using this feedforward patent. Of course the original patent has long since expired, and maybe I have missed some variation not covered by Quad to justify a new patent, but Quad's clever use of the output inductor as part of the feedforward network appears to have been abandoned, so how this could be an improvement is not clear. An obvious change is the name, from 'current dumping' to 'THX AAA', which may be a good move for the marketing department. Anyway, the new amplifier appears to be reasonably good from the rather limited specifications available, and is probably better than many other commercial designs. There are of course a few good DIY designs with comparable performance. The most impressive specification is for noise, but then looking further the closed-loop gain is unusually low, requiring 10V input rather than 1V in many designs including my own, so this alone gives a 20dB better figure for signal to noise ratio. There is fortunately an alternative switched higher gain level for more typical signal sources.
Incidentally, I mentioned a need for improvement to the Quad circuit to avoid distortion added by the nonlinear input impedance of the output stage loading the error amplifier. One solution I said was published in Wireless World, and I have now found the page, May 1979, page 71 in a 'Letter to the Editor' from Tore Hevreng. My own 'solution' is also mentioned in the 08-Jan 2014 piece.
Update: There are more detailed measurements of the Benchmark amplifier now published by Stereophile. There are some inconsistencies, for example the distortion at 1kHz at 15.4V has very different values in the distortion vs output level and the distortion vs frequency plots. One includes noise, but that should not cause a 4 to 1 increase. Also switching from maximum to minimum gain is said to only change signal to noise by 3dB, which seems 10dB less than we might expect. Anyway, even assuming the worst, the distortion results are excellent, possibly a little better than my MJR7 at high frequencies. Using dual die mosfets to match the power rating should also bring the distortion down to a similar level. Matching the noise would be impossible with my inverting circuit unless I reduced the input impedance to 1k, the input resistor adds 2uV noise. Using the same 2V input reference that gives a maximum possible 120dB signal to noise compared to 128dB measured for the Benchmark.
I mentioned on this page (01-Nov-2013) the difficulty I was having trying to determine how the output impedance of a common-emitter stage depends on the source impedance connected to the input. I found a published formula which gave obviously wrong results, including a negative resistance, and an IEEE paper claimed measured results entirely different. I have now found a piece in an old issue of Wireless World, Jan 1965, page 42, in a reply to a 'Letter to the Editor' from G.P.Hobbs saying that his own measurements gave a ratio of output impedances about 2 or 3 for source impedance varied from zero to infinity, although a calculation gave a ratio of 18. He suggested the problem could be that the 'typical' parameters in a data sheet are each a mean value for a large number of samples, but these values are not necessarily a consistent set. In other words no individual transistor could have exactly all the typical values. Although I don't find that entirely convincing, I have no better explanation, but anyway it is interesting that the confusing results I noticed were known and being discussed as far back as 1965.
Update: The h-parameters are certainly not independent variables, the input impedance hie is proportional to the current gain hfe, and also it is found that samples with higher hfe also have higher hoe and hre, so there is certainly some restriction to what is a consistent set of parameters. For an example see 2N4401 data, Fig11 to Fig14. The parameters are shown for a high gain and low gain sample. (Two curves in each fig. are not explained, on older versions of the data sheet these are shown to be the 2N4400, and again the higher gain version has all other parameters higher.)
The references on the feedforward amplifier page I just added appear to give conflicting advice concerning the best place to attach a temperature sensor to a power transistor. One says the heatsink close to the transistor base is best, another says the top of the transistor case. Both mention measurements being made, so who is right? One possible problem is that it depends on how the measurement is made, if we use a low heat capacity probe we may get one result, but if we use a higher capacity device, for example bolting on a TO-126 sensor transistor, then the low thermal capacity of the plastic case and high thermal resistance from case to junction will have a greater effect. For my own application with output triples there may be no need for a heatsink for the TO-126 transistors, and maybe the best idea is to clamp the two output stage transistors either side of the sensor transistor. Anyway, I need to do my own measurements, and add a warning that the references may not be as helpful as I originally thought.
Another point is that I wrote about bias adjustment for minimum crossover distortion in a conventional amplifier, and was foolish enough to say 'of course the conditions for minimum distortion were known before 1972'. I thought I should probably support that with a reference and started searching, but to my surprise found almost nothing. Eventually I found one example from 1969, a 'Letter to the Editor' from Peter Baxandall about designing quasi-complementary outputs , Wireless World, Sept 1969, page 416, where he writes about an output stage with 0R5 emitter resistors:
"When the output transistor current has risen to about 50mA, the reciprocal of its mutual conductance is about 0.5 ohm and the slope of the pair then reaches about half its final value of 2 A/V. Two pairs of this type (requiring complementary power transistors) would thus have an optimum quiescent current, for minimum crossover distortion, of about 50mA in each power transistor."
The same 0.5 ohm resistors and quiescent current 50mA were used in a design from 1961,'Transistor Audio Amplifier' by R.C.Bowes, Wireless World July 1961 Page 342 but without any explanation of how the correct current was determined, and I am not entirely certain it is still correct for that circuit. At the end of the article the author thanks P.J.Baxandall 'for many helpful discussions', which may be the explanation. That design used OC22 germanium power transistors driven from a transformer. The formula gm=Ic/Vt gives the same result for silicon or germanium, so the conditions for minimum crossover distortion are the same, at least for idealised devices with no additional internal base or emitter resistance.
A few older pages on my old website no longer have links to them, and some are really not worth keeping, but I found one example of a discrete transistor version of my original feedforward output stage amplifier. I may try building it sometime, but for now I have just corrected a few errors and repaired a broken link, and left it as 'work in progress' with the title Class-B Feedforward Amplifier.
The Guitar Pre-amp Mk3 is finished and works quite well. I have added a layout diagram, board design and photo. It needed a few alterations and so looks a little untidy, but apart from that everything works as intended. One important finding was that the J112 jfet in the tremolo circuit may not be an ideal choice. I originally used a surface-mount type PMBFJ112,215 made by NXP and that worked well after adjustment of the 1M preset for best effect. Then I tries a TO-92 case version of the J112 from Fairchild, and only a small effect was possible, and that needed the preset turned down close to the -4.5V supply. The specification for VGS(off) is -1V to -5V, so maybe the Fairchild version is closer to the -5V limit. Maybe the J113 is a better option with its -3V maximum specification. Anyway, I returned to another NXP device and again it works fine. Anyone wanting to try this circuit should maybe obtain both J112 and J113 to try.
I have updated most of the dead links on the website, a few items have vanished entirely and may no longer be available anywhere.
Many old issues of Wireless World are now available online at American Radio History going back to 1913. I started reading the magazine around 1968, but gave up soon after it changed its name to Electronics World in 1996.
I have just read a post on diyAudio mentioning an old advert for the Quad 303 where a number, maybe 20 or 30, were connected in series and the distortion at the final output was still within the specification. I don't remember that ad and can't find it via Google, but Quad did do that sort of thing. Some of the other Quad adverts are shown at pink fish media.
I also checked the calculation I mentioned of capacitor distortion appearing at the amplifier output, based on Bateman's worst case polyester capacitor producing -90dB 3rd harmonic with 4Vrms across it, for a 1V input at 1kHz there is only 6mV across the input capacitor of the MJR7, and making reasonable assumptions about distortion level versus voltage I estimate a contribution to amplifier output distortion around a tenth of a nanovolt. The minimum audible level from my own speakers at the 3kHz 3rd harmonic frequency I found to be 300uV, so the capacitor distortion reaching the output is then theoretically 130dB below audibility.
I still need to build the final version of the guitar preamp, but apart from that I need to do some website maintenance, there are a few dead links scattered around. There are some subjects I could write theoretical articles about, but that seems increasingly pointless, the website statistics show it is the practical designs most visitors look at.
I may still try one of the direct coupled amplifier designs I mentioned earlier, even though as I said then, the only unique feature is the ability to drive DC into the speakers, which we then need to prevent with speaker protection circuits. A direct coupled input also has its dangers, and achieves little more than saving one capacitor. Anyone who seriously believes they can hear differences between capacitor types can of course use something with audiophile approval, but note that the MJR7 distortion measurements include the effect of the cheap polyester type I used, so it doesn't appear to be a problem. I would be surprised if it added more than a few nanovolts third harmonic at 1kHz at the output. (I worked it out somewhere based on measurements by Bateman). My advice remains 'small size and low leakage' and maybe increased value to 4u7.
I have now updated Peak Current Requirement of Speakers to include links to the Google plots in the references at the end of the item. There is also a link to the Stereophile article which describes failed attempts to observe unexpected current peaks with music signals. That page of 'assorted items' was all written a long time ago, and there are a few dead links I need to update sometime.
I think I have made it all look more complex than necessary, the explanation of the current peaks in speakers caused by certain test signals is essentially trivial, at least if we assume linearity, and could be explained in one short paragraph:
Starting with a square wave voltage applied to a speaker, this consists of a fundamental frequency and a theoretically infinite series of odd harmonics. The resulting current has the same series of harmonics but because the speaker impedance has reactive components there is a frequency dependent phase shift of these harmonics, which changes both the shape of the current wave and its peak level. The square wave harmonics have the relative phases necessary to give the minimum possible peak amplitude, so almost any change from this gives a higher peak level, so the current wave will invariably have higher peak levels than if the speaker was a pure resistance which would have left the phases unchanged. There is no theoretical upper limit, shifting all phases by 90deg would give an infinite peak amplitude, limited in reality by the finite bandwidth and speakers having less than 90deg phase shift. This ignores changes in the magnitude of the speaker impedance as a function of frequency, a nominally 8R speaker typically falls to 5R or less at some frequencies, and this also increases the current.
The explanations given by other writers tend to concentrate on a more complex test signal cleverly chosen to maximise the current peaks with certain speakers, and their explanations involve 'back-emf', but what I wanted to demonstrate is that frequency dependent phase shift is a rather easier way to understand the effect, and applies for any rectangular test voltage, in all cases the phases of the frequency components start with the values needed to minimise peak level, so almost any phase changes cause increased peak level.
11-April-2015 (Updated 12 April)
Something I just found out is that you can use Google to plot a Fourier series. To see what I mean copy and paste the next line into the Google search bar and press return:
This is just the first few harmonics of a square wave. Google will plot it for you. Note that the peak amplitude is 1.18.
Try changing the phases in the above series, for example change the sines to cosines, equivalent to shifting them all by 90 degrees:
When this is plotted we find a very different looking wave and the peak amplitude is now 2.39, so increased by a factor of 2. If we had used the whole infinite series of harmonics instead of stopping at the 11th the peak level would have increased to infinity.
The reason I was looking at this was to add something to one of my articles to demonstrate that phase shifts can have a big effect on the peak level of a wave. A typical speaker has a frequency dependent phase difference between voltage and current, which is one reason why if we apply a square wave voltage the peak current can be higher than expected. Of course it works both ways, we could apply a square wave current to a speaker and find unexpected voltage peaks. This is all complicated by the impedance of a typical speaker varying in magnitude as well as phase as a function of frequency. My own speakers fall to 5.5ohms at low frequencies. I will try to find a link to an article I read in Stereophile describing failed attempts to find high current peaks in speakers using music signals. My own tests also failed to find anything to worry about. It appears to be one of those 'theoretical effects' having little practical importance.
Guitar Pre-amp Mk3 is the current version. There may be further changes before I am entirely happy with it, and I have not tried the input stage as shown, I am still using an earlier version, so that is the only part where component values are not necessarily final.
The Mk3 version of the guitar pre-amp is built and tested, and finally it is working more or less as intended. The opto-coupled light dependent resistor (LDR) is probably too fast, one with a slower decay time should remove some slight distortion noticeable at the bass end, so I will get another type to try.
The obvious approach is to add a capacitor after the level detection rectifier stage, but doing this caused serious distortion or instability, whether 2u2 or 1n was tried. Maybe I am just using too much gain round the loop at maximum control setting, but anyway leaving out the capacitor and relying on the slow LDR for loop stability seems to work quite well. The LDR specified for examples on the Elliot Sound Products page has a decay time of 1.5sec which I would have guessed was far too slow, but clearly it has been found to work, so I will try some slower LDRs.
The idea to use a jfet as the gain control in the guitar amplifier sustain function was motivated by the low current requirement, but I have also looked at the possibility of using an opto-coupler with LDR output.
I have a Silonex NSL-32SR3, bought from Farnell (UK), which should work ok at LED current under 1mA. Using this the number of dual op-amps could be reduced to two, though it may then be necessary to leave out the bass control. I need to do some measurements first to find out why the jfet version sounds more distorted than I expected. The latest but not final circuit is shown here as Guitar Pre-amp Mk2.
Much of what I know about physics was learnt from the Feynman Lectures. These are now available free online at:
The Feynman Lectures on Physics
These were written around 1963, so some of the material is out of date, but on the plus side many errors in the original books have now been corrected.
The next stage in the guitar preamp project is nearing a final version. I decided to include a sustain function, this is fairly simple to make, using a level detector to control a jfet used as a variable resistor. Jfets are not very linear in this sort of circuit, but adding half the drain signal to the gate effectively linearises it. Whether we actually want linearity is a different matter, it may sound better with some added 2nd harmonic. I also have included a tremolo circuit, with adjustable frequency and modulation depth. Again there is some uncertainty what will sound best, using a sine, triangle or square wave modulation.
I still want to use a single 9V battery, so this sets limits to what can be done. I need to use 3 dual op-amps, so current drain becomes a bigger problem. There are plenty of lower current types than the OPA2134 used in the first version but there are other specifications we need to worry about such as common-mode input voltage range, which can become a problem at low battery voltage.
I updated the page about cable skin depth to include the points made earlier on this 'latest news' page. One problem has been pointed out to me, that I said the surface field is determined by the instantaneous total current through the cable, which implies faster than light communication, how else can the current at the centre of the wire have any effect at the surface without a time delay. Given the low speed of electromagnetic energy inside the conductor the delay could be expected to be significant. I added a few words to try to explain that also, but I need to think more about that. More generally one of Maxwell's equations gives this 'instantaneous' effect, this relates the line integral of the magnetic field round a loop to the total current through the loop. The integral round a loop is equal to the sum of the integrals round two smaller loops or the sum for any number of smaller loops we choose to divide the original loop into. For any two adjoining loops the common boundary is counted twice in opposite directions in adding the integrals, so these all cancel and only the single outer loop counts, and includes the effects of all the inner areas. I also need to add something about the other term in the Maxwell equation which is the rate of change of the electric field, which is sometimes confusingly referred to as 'displacement current'. If the current at some depth in the wire changes then there must be a changing electric field, so this term needs to be included if we want to understand more completely how the surface field is related to a changing internal current.
I mentioned one of my recent projects was a guitar preamp, and I have added a page showing how this is progressing. So far it is very basic, just an experiment to see what characteristics are most useful. The idea to include asymmetric clipping was not found to add anything very dramatic, and has been abandoned for now.
I have been trying to understand US Patent 8466744 which lists my MJR7 amplifier under 'Other references'. The circuit shown as Fig.20 appears to be based on my old MJR6 circuit with just a conventional differential input stage added, and some component values changed. The claimed distortion figures, -167dB at 1kHz and -137dB at 20kHz I am almost certain are impossible for this circuit, unless I am missing something important, or unless stability is highly conditional, and anyway it appears the distortion figures are only simulation results.
I have now transferred the MJR9 feedforward amplifier page to the new website, and have rewritten it to include information about how it works and a simulation result, originally on other pages.
I have been asked whether my phono preamp circuit can be easily modified to work with moving coil cartridges. The 100R gate input resistor could be omitted, but the source resistor still adds too much noise. Just reducing the source resistor to something like 10R may reduce the noise far enough for all but the lowest output cartridges, but there are then other problems. The DC output voltage stability is not great to start with, and reducing the resistor will make that worse. The capacitor coupling to the next stage makes the DC level less important, but we would not want it to drift much more than a volt. Maybe a bigger problem is the drop in loop gain and consequent reduction in RIAA equalisation accuracy. The 68p compensation capacitor can be reduced to 10p, which should help a little. A further problem is that with the 10R source resistor the BF862 could have drain current up to 20mA, and this is greater than the minimum Idss of 12mA for the BF245C cascode fet. A different type with higher minimum Idss may be needed. Keeping the original source resistor and adding the 10R with a 2200uF series capacitor from source to earth could avoid some of the problems, but then some people will worry about having electrolytics in the signal path. With the 10R source resistor the 22R in the feedback network needs to be increased to 91R. Keeping the jfet current low however will reduce gm and increase its voltage noise, so that also needs to be considered.
My answer then is yes, the circuit can be modifed for use with moving coils, but it is far from ideal for that purpose, and I think it would probably be better to add another gain stage at the input or use a transformer.
I try to suggest alternative transistor types for my designs, but some of those I have mentioned so far are listed as 'obsolete' and likely to become unobtainable.
There are still a few types widely available which could work well enough in the MJR7 amplifier, the input stage can use npn type BC337-40, which has the advantage of a fairly low rbb' (30R), or the BC550C which may have a higher current gain. (High rbb' types such as the MPSA18 are probably not good for loop stability), and the BC560C pnp can be replaced by any similar high gain high fT type such as the BC557C. Gain over 500 and fT over 200MHz are the main requirements for the pnp, and those made by ON-Semi sometimes have higher specified fT than others, e.g. from Fairchild. The BC560C is preferable if still available, the noise contribution will be slightly lower, and current gain is usually over 550 at 5mA.
The KSC3503D and KSA1381E are still available from Mouser (US) and the lateral mosfets from class-d, but they have no gate protection zeners built in, so care is needed to avoid damage from static prior to connection in the circuit.
The guitar pre-amp I mentioned earlier has been built and tested, and is working well, but maybe can be improved. The adjustable asymmetric clipping turned out to have less audible effect than expected compared to symmetric clipping, although it is confirmed to be working as intended by checking with an oscilloscope. A simple tone control appears to be more useful in practice, and I have added an adjustable low-pass filter at the output, which also works as a volume control if the output feeds a low impedance such as headphones. I may add the circuit diagram here when I arrive at a final version.
Audio design would have far less point without the creation of music by musicians and composers, and we can all benefit from the encouragement of new talent. I mentioned that my son is learning to play the electric guitar, but he is also a composer of synthesised electronic music. Some of his work can be found at Bandcamp. It isn't the sort of music I usually listen to, so it's hard for me to judge, but I quite like the second track. Have a listen and see what you think. I have to confess I do sometimes make suggestions, such as 'more dynamic range', 'less repetition', or 'what it really needs is some tubular bells.'
The tracks at Bandcamp can also be downloaded in various formats for a fee, but you can contribute as much or little as you like, anything from zero upwards. Bandcamp keep 15% and pass the rest on to the artists.
Searching for 'audio amplifier design' via Google today had my own website in third place on the first page. The only problem is that it is my old site on Angelfire, which has annoying pop-up advertising, my new website is somewhere on page 5.
In second place was the wikipedia page about audio power amplifiers which looks about 40 years out of date, including uncritical reference to the work of Otala on 'TIM' (with no mention of earlier publication by Daugherty and Greiner). It has been pointed out often enough that what matters is the magnitude of the overshoot (in volts) not the percentage relative to its final level.
I was pleased to see at the top of the list was the Rod Elliott website. It has lots of useful information for anyone interested in amplifier design. My own website is intentionally limited to topics I think are not covered very well elsewhere, and slightly unconventional designs, so more useful as 'further reading'.
Anyway, I am tempted to delete the new website, or maybe replace it with something completely different, and leave the old 'free' website as a possibly more permanent record.
I mentioned that I have not built anything for a long while, but I do have a project or two. My son is learning to play the electric guitar, so I wanted to make a suitable pre-amp with a few 'effects'. Reading about guitar amps is something of a revelation, the usual hi-fi aim of low distortion becomes relatively unimportant, with the amplifier intended to contribute to the final sound. The pickup output is higher than I expected, and so is the recommended load impedance of over 500k. A unity gain input buffer with a jfet input op-amp such as the OPA2134 looks like a good place to start, plus a second stage op-amp providing frequency response equalisation and adjustable clipping. Apparently germanium diode clippers are favoured by some, and I have a few OA5 to try. Maybe optional asymmetric clipping to add some even order harmonics is good also.
I left one problem unsolved regarding my phono preamp. I originally included 100k offset adjustment preset pots for the output stage op-amps, but found these difficult to adjust. For my own use that was of no importance, my main preamp has capacitors feeding the volume control, but for anyone needing low DC at the phono preamp output I should suggest a solution. The cause of the problem is the interaction between channels via the 5k6 resistor (R10) forming part of the 'rumble filter'. The procedure I followed of just trying to set one control and then the other is not guaranteed to work. A better idea is to short one op-amp inverting input to earth and then adjust the other channel preset for zero DC output. Then remove the short and set that channel also. That way both op-amps have their inverting inputs set close to zero, and the outputs are also both close to zero. I tried this, and could easily get both outputs down to 0.2mV or less, compared to typically 2mV without the adjustments.
Looking again at the animated plot at Some Skin Effect Notes the points on the plots at the left hand side oscillating between +1 and -1 represent the field levels at the surface and outside the conductor, and that field may travel close to light-speed carrying energy to the load at the end of the cable, while the internal variations travel into the wire at much lower speed, e.g. 3m/sec for a 60Hz signal. I suggested it can easily be understood from the animated plot why the delayed and reversed direction currents at different levels in the wire are not a problem.
The total current can be found by integrating the current density through a cross-section, and the time delays and current reversals may seem to complicate this, but one of Maxwell's equations, for the curl of the magnetic field H, tells us that the line integral of H round the circumference of the wire is proportional to the instantaneous total current along the wire, so the value of H at the surface tells us that the total current including delays and reversals is proportional to the surface current with just the 45 degree phase difference between J and H at the surface mentioned previously. The animation assumes a wire of large diameter compared to the skin depth, which is rarely the case in audio frequency applications, so the less delayed current near the left side of the plot is all we are concerned about and the resulting phase shift may be far less than 45 deg.
There are other ways of looking at this. The 60Hz example has a time period of 16.7msec, enough time for the internal field to travel 50mm into the wire, if it was thick enough. Using a more typical 1mm diameter wire the 'slow' internal field will reach the centre after 167usec, at which time the surface 60Hz wave has only changed phase by 3.6 degrees. At higher frequencies the refractive index is lower and the internal field travels faster, so it gets to the centre of the wire quicker, but the phase of the surface wave has changed more. Anyway the internal field is not some highly delayed version of the signal, it is just phase shifted a little, with higher phase shift at higher frequency. The internal energy storage with phase shift increasing with frequency is equivalent to just an internal inductance, in effect in series with the external inductance, though not a simple fixed inductance, its value changes with frequency. The reducing skin depth at high frequencies means the internal energy starts to drop when the skin depth becomes comparable to the radius of the wire, so the internal inductance falls. A good account of this internal inductance can be found at Electromagnetic Waves In Matter.
A possibly important question is whether the added phase shift is a linear function of frequency so that there is a constant time delay. An interesting link covering this question is Effects of wire diameter and spacing where Fig 14 shows that group delay is virtually flat up to 25kHz for 1mm dia wires, but is not so flat for 2mm or greater, but even then the change in delay at 25kHz is mostly under 50nsec, so fairly harmless. The results shown there are for a 3m length of cable driving a 8R load.
I may eventually transfer these pieces about skin depth to the relevant page, but I will try to improve the explanations a little, I am not sure I explained it clearly. I maybe need to explain why the internal inductance is in effect in series with the external inductance, otherwise we could imagine the internal and external fields are two separate signals travelling along the cable. We could say the internal field is caused by the external field, but there are perhaps equally good reasons to say the external field is caused by the electrons in the wire, so maybe causality arguments are not helpful. Either way the two signals are linked and must travel along the cable at the same speed. The electrons in the wire need not travel along at the speed of the external field, it is only changes in electron density which match the speed of the field.
Also to be explained is how the field travelling into the metal is related to the field travelling along the wire, I wrote something about that in the transmission line page, which could also be improved, that was originally an unpublished 'letter to the editor' about an article in Wireless World many years ago, so it is probably trying to answer questions no longer being asked by anyone.
I maybe need to mention that exact solutions of this sort of problem are often extremely difficult, and explanations of the sort I sometimes attempt are usually over-simplistic and only close to the truth under a limited range of conditions, so it may in practice be easier to just measure what happens under the conditions of interest.
Looking back through this page I see I have not actually made anything since October 2012. Part of the reason is that I have my own audio system as good as I need it to be. My hearing has deteriorated to the point where I am unlikely to hear any further improvement, and the only worthwhile changes left are the speakers and listening room, which I hope to improve soon. Much of my listening is done using headphones (Sennheiser HD238) via my computer soundcard, which my measurements confirm is far better than my CD players.
I wrote a piece some time ago about conduction in metals, but realise some of that is at least simplified if not actually wrong. The part about skin depth certainly needs improving. It is based on the explanation in Feynman's Lectures Vol.2 which involves the refractive index of the metal. I learnt all that over 40 years ago and never thought much more about it since. There is an explanation in Wikepedia in terms of circulating eddy currents cancelling the current at the centre of a wire, but I think that is not the best way to understand it.
I only recently learnt that the skin depth is just the wavelength of the signal divided by 2 pi. The relevant wavelength is inside the conductor, and for example in copper at 60Hz the skin depth is 8mm and the wavelength is just 5cm. The wavelength in a vacuum is 5000km, and the ratio of wavelengths tells us the refractive index of copper at 60Hz is around 100 million. The velocity of the electromagnetic field into the interior of the conductor is therefore a surprisingly slow 3m/sec. One result of this is that the surface current can change polarity while the current further below the surface is still in the original direction. In other words the current can at times be in different directions at different depths. A good animated plot can be found at Some Skin Effect Notes (But there are errors in some of the equations.) Note that the magnetic field H lags the current density J by 45 degrees. Note also that the animation looks the same for a wide range of frequencies because of the fixed ratio between skin depth and wavelength, and because the horizontal axis is in multiples of skin depth rather than actual distance.
It may be tempting to think this must cause problems in audio cables, if part of the current is determined by what happened maybe as much as a msec earlier would that not 'smear transients'? I think that was actually suggested in a published article some years ago. The reason why it is not a problem I am sure has been explained somewhere. I think it should be easy to explain by reference to the animated plot, but I'm a slow thinker, so I'll get back to that in a while.
That reminds me of a story about a professor telling some theory to his students and saying 'I'm sure this is obvious, just give me a minute to work out why'. Then after 3 hours deep in thought, and long after all his students had left, he announced 'Yes! I was right, it really is obvious!'
The phono pre-amp pages continue to be the most popular. This is closely followed by my CD player modification page, which is perhaps unfortunate because the conclusion was that none of the 'improvements' tried actually made any measurable difference.
I have added another rumble filter design to the references at the end of the Phono Pre-amp Design Theory page. This version is from Dimitri Danyuk, and is a second order version which includes a high-pass filter which can reduce horizontal rumble effects in addition to the cancellation of vertical rumble. The two channel filter could be adjusted by a single variable resistor, which at first I thought was not possible, but I now see it really can work.
On this page a while back I said "For a given speaker almost any cable will add some frequency dependent amplitude and phase variations". There are two types of distortion, linear and non-linear. The non-linear variety adds new frequencies such as intermodulation products, and can be detected with the usual distortion measuring equipment, and if it can be detected at all for a copper cable it is invariably found to be down near the limits of measurement (For example see Cable distortion and dielectric biasing debunked by Bruno Putzeys.)
Various mechanisms have been proposed for cable nonlinearity, ranging from unlikely 'micro-diodes' to known effects such as magnetoresistance. My own view is that there is no point looking for or inventing different explanations unless some reliable measurements can provide data sufficient to distinguish between them. Even for the known effects it is difficult to find any convincing calculations of the distortion levels to be expected in typical copper speaker cables.
Linear distortion adds no new frequencies, so each frequency component can only be changed in level or phase, so frequency dependent amplitude and phase variations are the only effect possible.
Some years ago I wrote a piece in my physics section, Cable Impedance, showing how even the worst possible example, with a current step driving a lossless open ended line, which produces a continuously repeated upward step output having no resemblence to the single input step, is just the same effect as a single capacitor for band-limited audio signals. I recently noticed that AIM-Spice includes a lossless transmission line model, so I have extended that page to include a few more examples, together with amplitude and phase plots to show that nothing bad happens within the audio range.
Maybe sometime I will extend the article further to include resistive losses and properties of dielectrics, but I am fairly sure nothing important would be revealed. I will continue to use my cheap zip-cable, but of course anyone using speakers with extreme low impedance dips or amplifiers with stability problems may need to select cables more carefully.
The idea for the MJR11 power amplifier started from the phono-preamp input stage. The simulations to investigate stability revealed that the 2nd stage transistor is more critical than I expected, a type with low base-spreading resistance rbb' gives a better stability margin. I suggested a BC560C would be good enough for the preamp, and checking again shows the stability margin is just adequate, but substituting a BC327-40 with rbb' 30R is a big improvement. My own preamp uses the 2SA1085E with even lower rbb', but these are unobtainable from most reputable sources. Few data sheets specify rbb', and some Spice models appear to be unreliable, so some care is needed when specifying alternative types not actually tried.
I mentioned a design for a higher power direct coupled amplifier, and although this may never get made here is a preliminary example of the sort of thing I was thinking of, called the MJR11-Mk1. So far it has a few problems which need sorting out, but I like the general idea.
Looking at the website statistics the most popular page is my phono pre-amp design. That is certainly far more complex than is really necessary, but none of the components are expensive, so I just made it as good as I could without going to ridiculous extremes. I should maybe add an optional input stage for moving coil cartridges. There are plenty of published examples of MC input stages, but if I think of anything different I will include it sometime.
Thinking about new projects for the future, power amplifiers are not high on the list, the excellent results from the MJR7 would be difficult to improve on. One option would be a direct coupled higher power version. Dual or triple mosfets will have lower open-loop output impedance, which will reduce the effect of reactive loads, and so allow higher feedback, and should also have lower open-loop distortion. Achieving low distortion with higher power amplifiers is therefore less of a challenge, but lower distortion with a single pair of mosfets is also easy enough using feedforward as in the MJR9.
Just using dual die mosfets in the MJR7 and increasing the supply voltage should be enough to push the power output well beyond 100W. I have seen amplifier modules claiming power rating up to 200W with just single lateral mosfets, which may be ok with resistive loads, but reactive speaker loads with impedance dips in the audio range are a different matter. I usually assume a minimum impedance of 3R but a few speakers fall even lower.
I am not enthusiastic about direct coupling, it just adds more problems. Its only unique achievement is the ability to drive DC into the speakers, which we then need to prevent with speaker protection circuits. The avoidance of amplitude and phase errors at low frequencies can also be a feature of direct coupling, but my usual approach when using capacitor coupling is to set the -3dB frequency to about a tenth that of the speakers so that additional amplitude and phase errors are relatively insignificant.
The entry for 14 June 2013 mentioned "The original plan when I set up this new website was to add some products for sale, maybe a few different kits, some hard to find transistor types and so on." That is still a possibility, but past experience suggests my existing designs are unlikely to attract many buyers, so maybe the 'higher power direct coupled amplifier' could be worth working on, I have a basic idea using a single jfet input something like the phono preamp input but with servo control. That would also need to include reliable speaker protection, I would hate to be blamed for destroying anyone's speakers.
If I wanted to be cynical I could always produce some 'exciting new concept' in speaker cables and sell them at an astronomical price. Such products continue to be available, with impressively imaginative advertising, so there must be plenty of potential buyers around. My own speaker cables are an unknown brand, they look like fairly cheap 'zip-cord', which came with the speakers when I bought them secondhand, and I never found any reason to change them. I accept that for a given speaker almost any cable will add some frequency dependent amplitude and phase variations which in extreme cases could be audible, but anything beyond that I suspect is 'marketing'. Some of the more expensive cables probably are 'extreme cases' and really do sound different, or in some cases cause instability in badly designed amplifiers. (My original MJR7 was an example of this, but fortunately one of the early constructors discovered the problem and kindly told me about it, so a hasty modification was needed, which is why I now use a RC Zobel in the 'wrong' place, which turned out to be the right place.)
One more point regarding the 6th edition of 'Audio Power Amplifier Design', on page 333, and in Fig.13.5(b) shunt compensation is criticised on the grounds that for the example given a shunt capacitance of 44nF is needed, together with 155mA from the VAS to drive it. This appears to be another example where what may be true for some BJT amplifiers with moderate feedback is not necessarily true for mosfet designs with high feedback, my own MJR7 being an example needing only 100pF driven by a stage running at 5mA. Shunt compensation may be less problematic for what is in effect a two stage circuit.
One possible advantage to shunt compensation is that it is minimum phase, unlike the Miller compensation in which there is feedforward through the capacitor adding more phase lag. This is often dismissed on the grounds that the effect only becomes significant far beyond typical loop unity gain frequencies. This is usually true, but only for a stage with high gm, which is not necessarily true close to clipping where the gm of the VAS falls. For Miller compensation the phase shift may therefore increase near clipping, but for shunt compensation if the impedance at the VAS output falls near clipping, as it does in my MJR7 circuit, then phase shift added by the capacitor is reduced, and stability margins get better instead of worse.
I have never tried comparing the clipping behaviour with reactive loads for the two compensation methods, so I don't know for certain how important it is in practice. The feedforward problem was mentioned by Baxandall in his 1978 Wireless world series (part 3), but only his remark in part 4 about shunt compensation being 'in all respects sub-optimum' seems to be remembered, even though he pointed out that just adding a series resistor could solve the most serious problem.
I have been looking at a US patent for a 'new' feedforward amplifier, No.8,004,355. Maybe I am missing something obvious, but this looks identical to the Quad current dumping circuit from 1971, apart from a few changes to the bridge components which appear to lose one advantage of the Quad circuit, that the output inductor needed for stability into capacitive loads is part of the distortion nulling circuit. My own MJR8 and MJR9 from 2007 were also based on the Quad idea, as explained on the MJR8 page. The original Quad patent has long ago expired.
The only real problem with the Quad circuit is that the nonlinear input impedance of the output stage loads the driver stage, and any resulting distortion is not nulled by the feedforward. There are a few solutions, including one published in the letters pages of Wireless World, and also one of my own untried ideas (Output Stage Variations, 2006) where the output stage drive is taken from a driver stage supply line rather than from its output. The distortion figures for the Quad 405 were nothing special, and my own experiments with feedforward were mostly discouraging, it was easier to achieve low distortion using conventional negative feedback. The point of the Quad circuit was to achieve adequately low distortion without any adjustments or accurate control of quiescent current, which has considerable advantage for a mass-produced commercial product.
I have been looking at the latest (6th) edition of 'Audio Power Amplifier Design Handbook' by Douglas Self. Some sections can be read for free on Google Books
There are inevitably some points I disagree with, but otherwise it looks good. The book concentrates almost entirely on BJT amplifiers, so some of the conclusions are not necessarily applicable to mosfet designs. Here are a few points worth mentioning:
On Zobel Networks he writes about the usual RC Zobel that it is 'always fitted on the inside (i.e. upstream) of the output inductor.' Also the resistor value he says 'approximates to the expected load, and is usually between 4R7 and 10R'. In my MJR7 I used a 1R resistor plus 100nF capacitor, and added the RC after the inductor. At very high frequencies the resulting load is still mostly resistive, about 2R5 total. I explained somewhere why I put the RC after the inductor, it concerns stability with capacitive loads around 2nF which resonate with the inductor close to the unity gain frequency. It may be that BJT output stages are affected less by such problems than those using mosfets because of their usually lower open-loop output impedance.
In Fig 14.11 the 'optimal coil shape' for the output inductor shows that my own coil design is hopelessly inefficient, it should have higher area and shorter length. What I didn't find mentioned there is another factor which is that increasing the area increases the pickup of interference from varying magnetic fields, although the problem of pickup from external fields is mentioned in the section on 'Coil Placement Issues'. Higher area gives higher pickup, but the lower number of turns for the same inductance reduces pickup. My calculation suggests the 'optimal' coil is worse by a factor of 1.7, but this is probably not very accurate.
On page 108 it is stated that there are only two cures for output capacitor distortion, both involving expensive capacitor types. This is puzzling because a third method is mentioned later, where the capacitor is included at least partly inside the feedback loop. This approach was probably first used in a famous design published in 1956 by H.C.Lin., the circuit of which appears in chapter 30, Fig.30.5, followed a few pages later by an example by Hardcastle and Lane in Fig.30.8. It seems unlikely that those designers were worried about reducing capacitor distortion, but inclusion in the feedback loop can be highly effective for that purpose.
I mentioned a while ago that I was working on a 'beginner's guide' about amplifier design. That never got finished, and the only section I did finish is really not very good. I think the problem is that I never studied electronics at an elementary level, except as part of a physics course, so I have no good ideas where to begin or what to include. Starting from Feynman's Lectures Vol.2 is probably not good advice. Even so, I have added a page called Transistor Amplifier Design for Beginners which just covers a few details I remember having difficulty with many years ago. Beginners now inevitably have a different starting point, there is almost unlimited information available on the internet, compared to when I started learning, at which time Wireless World was almost the only easily available source.
I also wrote a page about the common-emitter output impedance, trying to explain how it depends on the source impedance. I finally realised I don't fully understand that topic, and I need to do further investigation before either adding the page or admitting defeat.
I have rebuilt my old Angelfire website so that it includes most of the material on the new website, but continues to include some of the older pages which were never transferred in an Archive page. One of the pages listed is the MJR9 feedforward mosfet power amplifier. Unfortunately there is a limit of 20Mb on the 'free' site, so something had to be left out, and that was the 'Constructor's Page'. (Update June 2014, I have added a reduced version of the Constructor's page to the Angelfire site, leaving out measurements, which mostly only applied to earlier versions). As far as I can see the free websites have no time limit, I just found an even older website I made over 12 years ago which I never update, but still it continues to exist, and I like the idea that my website at least could go on into the indefinite future.
The common-emitter output impedance Zo is actually a more difficult problem than I thought, unless of course we just take it to be 1/hoe and look up the h-parameters in the data sheet. Many data sheets fail to include even that, but if we do get the value there is still the question of how the output impedance is affected by the source impedance Zs. I have only a vague idea about this, so I started with a Spice simulation, which showed no effect from Zs. Spice models are not necessarily reliable, so I searched for a formula, and found one in Small Signal BJT Amplifiers (dead link) section 3.6 starting on page 19. The formula there is Zo = 1 / (hoe - (hfe.hre / (Rs + hie))) . Taking the h-parameter typical values from an old BC560B data sheet I got Zo = 100k for a very high Rs, and for zero Rs I got Zo = - 59k. Yes, a negative resistance. Trying again with a more recent data sheet for the 2SC4117 gave more reasonable results 400k and 2.8M with high and low Rs respectively. To make matters worse I found an IEEE paper Output resistance of the common-emitter amplifier which states that the ratio of Zo values for high and low source resistance is 1.5, and shows measurement results which appear to support this. Worse still, the higher Zo is for higher Rs, the opposite of the formula, and not what I would expect. The article does agree with my own observation that Spice simulations can fail to show any effect from Zs.
If we just use Zo = 1/hoe we may look at the data sheet for hoe and for the BC560B find its typical value to be 10uS, but we also find a maximum value 60uS. The ratio of maximum to typical is 6, far greater than the ratio of maximum to typical current gain hfe, which is just 1.5. The important thing to learn from this is that it is a bad idea to design circuits where either hoe or hfe are critical, which would require testing and selecting transistors. If we want very high output impedance there are more predictable ways to do that, such as adding a cascode (common base) stage.
I promised a long time ago to add some pages for beginners, and I have been trying to write a page called 'Transistor Amplifier Design for Beginners'. The trouble is that to make it easily understandable it would need to be simplified to the point where it is at least misleading, and in some ways just wrong. I don't know how to explain the output impedance of common-emitter stages in a simple way. Just drawing equivalent circuits such as the 'common-emitter hybrid pi-equivalent' gives no real understanding of what is happening or why. For example how can there be a resistor between collector and emitter when there is no direct connection between them, all current from collector to emitter must pass through the base region. I will carry on trying to get it right, but may have to change the title to 'Amplifier Design for Advanced Beginners'.
My search for good low noise transistors continues. The really hard to find types are those combining high current gain with low rbb', and so possibly having low current noise and voltage noise. The 2SC2547E I originally used is one of the best, with high gain and rbb' sometimes claimed to be around 2R. The 2SC2240BL is about 40R. The only good and available example I found so far is the 2SC3324BL, rbb' less than 20R, which is a surface mount type looking very similar to the 2SC2240BL, and possibly about to become available from Digi-Key, but with £12 delivery charge to UK unless you spend £50.
I was looking at the RS website, and noticed that the 2SC2240BL is now listed as 'discontinued', and apart from a few of uncertain origin avilable via eBay at well above the old RS price these now appear to be unobtainable. I still have some for my own future use, but I am looking for alternatives. The high gain MPSA18 are also listed as discontinued. Farnell may have them available soon, but only with a 2000 minimum order. They have a far higher rbb', around 800R, than the 2SC types I originally used, so no good for low impedance circuits such as mm-preamps, but for the MJR7 with over 10k series input resistance that is the least of our worries, and the high current gain is far more important. The high rbb' will however have other effects including adding phase shift round the feedback loop, so the MPSA18 may be a problem. There seem to be endless variations on the BC550C, (rbb' = 160R at 0.5mA) which may be good enough, but genuinely low noise types are becoming rare. Both input stage transistors need high current gain for best results, 500 or more is good, and if using alternative types or if they could be fakes the current gain should be checked. Even some very cheap multimeters have a current gain measuring function. I will try to do some tests on some of the more easily available types to see if any can be recommended for the MJR7. Past experience suggests there can be considerable difference between the same type from different manufacturers.
I have added a page of links to interesting websites, including a few related to audio and electronics plus some other topics, from physics to food and drink.
The original plan when I set up this new website was to add some products for sale, maybe a few different kits, some hard to find transistor types and so on. I was expecting to 'retire' a few years early, so I would have lots of free time, and for financial reasons needed to be at least nominally self-employed. That may happen sooner or later anyway, but for now everything is held up by problems selling my apartment, the local housing market is slow or stopped. At some point I may copy the new website content onto the free Angelfire site so that there is some chance of it continuing to exist indefinitely.
Apart from that, no news. It's summer and I rarely do much electronics when the weather is good.
I have probably already written more than enough about amplifier design in general, and far too much about slew rate, but I was playing around with a simple simulation and found some interesting results which I had previously suspected but never checked, so just one more article, then I promise to never mention slew rate again. The point, if there is one, is to compare the relative importance of slew-rate and gain-bandwidth for the accurate amplification of transients. I already knew the answer, but I was still surprised at how little difference a higher slew-rate limit makes.
As soon as we introduce a band-limited input signal these considerations become entirely unimportant, and the accurate amplification of audio frequency transients is determined by the phase linearity in the audio range, which is why I showed the phase response from 1kHz to 20kHz for my MJR7 design, but rarely mention its slew rate limit. The article also shows that the low-pass filter used in the MJR7 is practically the only effect on step or square wave inputs. Anyway, here is Slew Rate Part 2.
I still have a page of 'untried and abandoned' designs on the old website, and having accumulated a few more I have included two of these here on a new page Mosfet Designs.. As before, although they may be of some interest and could possibly be made to work well enough with further development, these have been rejected for good reasons, and are not recommended in their present form. A few more examples may be added eventually.
I had a few enquiries whether I was going to make boards available for the phono pre-amp, and I did consider this possibility, but I have no great enthusiasm for making boards, and there are anyway board makers willing to make small quantities at a low cost from gif files similar to those I give, for example this one found on diyAudio. I think I would probably want to charge more if I made them myself.
There is another reason to be reluctant to sell boards or kits, which is that I only made one pre-amp for my own use, and I have no feedback from anyone trying the circuit. Anyone with sufficient test equipment and the knowledge to track down and solve problems will probably be able to make their own boards anyway, so I don't want to encourage less experienced constructors to make this circuit until I am more certain of its repeatable performance. It took until the Mk3 version of the MJR7 before I had enough feedback from constructors to persuade me that it had developed into a reliable and repeatable product, and I made a few boards. Even then I didn't promote it to any great extent, and suggested it was not suitable as a beginner's project.
Well, I still have no bright ideas for another electronics project. I have been working on an addition for my physics section, but that also is maybe going nowhere. I just found a musical link mentioning both the Higgs boson and Miley Cyrus, I'm not sure how they are connected, it's Higgs Boson Blues by Nick Cave. The sound quality is not great. It somehow reminded me of the Britney Spears' Guide to Semiconductor Physics which is actually a serious physics site. Also, I have been listening to 'Anywhere I Lay My Head', which is Scarlett Johansson singing Tom Waits songs, made even more remarkable by the appearance of David Bowie as a backing singer on some tracks, including Fannin Street.
Components for the MJR7 continues to be a problem, the vertical mosfets are still widely available, but very expensive in some parts of the world, with an inevitable consequence that fakes are also being sold. The versions supplied by class-d are maybe the lowest cost at £3 each, and if bought direct fairly certain to be genuine. The cost of postage and any customs/import charges may be discouraging for some locations.
My website provider makes available many statistics, for example the number of pages downloaded for different countries, and the most popular files. The top country as expected is USA, followed by UK and France, but with occasional surprises, at one time I was popular in Finland. The top links to my site include a few of the audio forums including audax.fr, audiokarma and diyaudio. The top file is currently the phono pre-amp page, but there is also some interest in the distortion testing files. The pdf document listed there was written 35 years ago, when I was a student and a relative beginner, so there are some things I would now do differently, but I don't think there is anything seriously wrong in that original version. The unity gain inverting amplifier used for some measurements could easily be improved, but was good enough for most applications back then.
I have been taking a break from electronics for a while. The signal generator designs I was experimenting with work well enough, but nothing exceptional, and I may try one further idea some time. My other great interest is physics, but I have too limited mathematical ability to make any worthwhile contribution to that subject, and just struggle to understand some of the current theories. This week I was reading about black hole 'information loss', and the firewall theory.
I have decided on a few practical circuits for my signal generator project. They are just interesting ideas I want to try rather than anything likely to have exceptional performance. For anyone wanting to find some better designs a good place to look is a diyAudio discussion, Low-distortion Audio-range Oscillator, now up to 83 pages and with lots of useful links. My own first effort using a twin-T oscillator was somewhat discouraging, but did show that the simple light bulb method of level stabilisation can still achieve low 3rd harmonic distortion, better than -120dB. The control loop adjustment was far too critical however.
I am starting work on my signal generator project, but it may be some time before I have anything to report. I have moved an earlier section from this page to a new Signal Generator Design, Part 2 page. So far it just includes a few design ideas, not anything final. I need to order a few parts for some initial experiments. My design aims include low cost and simplicity, not necessarily the ultimate low distortion, I will be happy with anything under -120dB. There are plenty of excellent published designs, including some with distortion under -140dB. Some of the best use Silonex optocouplers, such as the Silonex NSL-32SR3, available from Farnell (UK), so I will also order one of those to try.
(I checked the Silonex data sheet, and found that although based in Montreal they also have an address on Princes Street, Ulverston, in the English Lake District. I was there just a few weeks ago with my family, walking down Princes Street from the train station. Ulverston is most famous as the birthplace of Stan Laurel. I didn't notice Silonex, but we did find the statues of Laurel and Hardy outside the Coronation Hall theatre.)
There has been occasional disbelief that such a simple power amplifier circuit as the MJR7 could have very low distortion figures, and maybe my unconventional distortion measuring technique using signal nulling is partly to blame, so I am pleased to have been given a link to a page of measurements of the latest Mk5 version using more conventional test methods. The 1kHz distortion figures are even lower than my own tests suggest.
L'amplificateur de Mike Renardson à transistors Mosfet by Forr.
The article is in French, but for non-French-speakers most of the test results are still easily understood. A not entirely accurate translation can be obtained by copying and pasting sections of the text into Google Translate.
The tests include inductive and capacitive loads, revealing that there is no serious effect on distortion levels. I was surprised to see the test with a 16.8uF load, I never tried more than 4.4uF and had expected instability with higher values. Checking a simulation I find that the feedback loop phase shift does then go well past 180 deg. but the loop still has gain far enough above unity to remain at least conditionally stable.
Other tests include supply rejection, which again I am pleased to find are in agreement with my own observations. Many thanks to Forr for some impressive work.
I have added a few updates and new sections.
There are various design notes already scattered among the various MJR7 pages, but I have collected a lot of this onto one page, MJR7-Mk5 Design Notes which summarises the design features and its advantages.
I added a few diagrams to the end of the Capacitor Distortion page. They really add nothing new, but may help dispel unnecessary concern about choice of input coupling capacitors. My recommendation was, and remains, to use a low leakage type with small physical size to minimise interference pickup.
I continue to add and rewrite sections on the Assorted Items page. I use that page for more doubtful items with in some cases too much personal opinion, and 'facts' I am less certain about.
The only project I have been thinking about recently is a design for a low distortion signal generator. The one I use at present has about 0.007% distortion at 20kHz, and I made it around 1982. Even back then there were plenty of better designs around, the best I was aware of was published as 'Spot-frequency distortion meter', by John Linsley Hood, Wireless World, July 1979, pp.62-66, with distortion 0.00015% at 1kHz. This used the R24 glass encapsulated low power NTC thermistor for level stabilisation, and these are practically unobtainable now. An important specification is the 'dissipation factor' which is 0.02mW/deg.C for the R24. Available types such as the Epcos G540 series are 0.4mW/deg.C and so need to be operated at much higher power dissipation. Anyway, I never tried thermistor control so I probably don't understand all the problems and design requirements. My own design used a jfet for level control and switched ranges covering 10Hz to 100kHz, sinewave and squarewave. A more or less similar circuit was published by Bob Cordell in 1981, which I unfortunately had not seen, otherwise I would have realised my choice of op-amp and jfet was far from the best even then. I have added a page with some initial ideas.
I have updated and rewritten a few pages. One addition is to the MJR7 Constructor's Page immediately after my own version near the end of the page is a very impressive version of the Mk5 using a double-sided board and a highly original layout. My own design looks a little untidy in comparison, but the single-sided board should be easier to make for less experience constructors.
I have added a page showing my efforts to improve my Technics CD player, 'Technics SL-PG390 Mods'. None of the changes I tried produced any obvious measurable improvement, but at least my experiences may be a useful guide to what not to do. I was anyway already happy with the sound quality, so I will give up on any further attempts at improvement for now and just enjoy the music.
I originally found that the 1kHz at -60dB test track spectrum had higher components at the 50Hz supply frequency and its harmonics than before the modifications, so I thought I had made this worse, but what I forgot to check is the effect of reversing the two pin power connector, and having now checked this I find that it does make a difference to the measurement, and connected one way it gives results almost identical to that before the mods, so it may be that I had the connector reversed between the two measurements, so I am happy to believe my efforts really did no harm. The 100Hz component, which may come from the full-wave rectified supply rather than from the transformer field is actually reduced, so maybe the improved supply smoothing did have some benefit. Moving the output section further away from the transformer would almost certainly reduce the 50Hz and 150Hz components.
I now have a Technics SL-PG390 CD player to experiment with. Initial tests show that it is already quite good, but maybe it could benefit from an op-amp upgrade and some attention to supply regulation, and maybe replacement of the muting transistors with a relay. When I have tried a few modifications I will add a page to report my results.
I have also finished my own final version of the MJR7-Mk5 power amplifier, which I am now using in place of an older mosfet power amplifier I made over 30 years ago. I have added a few photos of my own version to the end of the 'Constructor's Page'. This uses a recycled case from the old amplifier, so it looks a little untidy, but works fine.
I have finished the phono-preamp, and it is in a separate screening box and connected to my 'new' pre-amp, which as mentioned earlier is built in an old Cambridge ATAC3 case. I kept the original tone controls apart from replacing the op-amp, there is a bypass switch to disable them but I can hear very little difference when the controls are in the flat position. I tried my old Technics EPC205C-Mk3 cartridge with my Pro-Ject 1 Xpression turntable, expecting the worn stylus to sound bad, but it actually sounds remarkably clear. I previously only used that cartridge with a Thorens TD125 and it may be that slightly different alignment means that a different area of the stylus is now contacting the records, but whatever the explanation it does sound good to me. The Technics cartridge is a low output type, about 0.5 mV/cm/sec, but even so the noise level from the pre-amp is low enough, far below typical record surface noise, so I have no plan to try the active input resistance idea to reduce noise further. The 'rumble filter' appears to be effective, listening on headphones where lack of acoustic cancellation could make rumble a problem I still found it reasonably unobtrusive.
I was thinking of upgrading some of my other signal sources. These are a Marantz CD273SE cd player, a Nakamichi CR-2E cassette recorder and a Denon TU-260L FM tuner. Listening and comparing these the one standing out as most in need of improvement is the Marantz cd player. I also have a Panasonic SL-CT700 portable cd player, and much prefer its sound. Looking for some technical information it appears that the Panasonic uses a MASH multi-stage noise shaping single-bit D/A converter as used in many Technics players. I had just assumed that this was a low cost inferior alternative to the more expensive multi-bit converters as used in the Marantz, but I see even some 'top of the range' players also use single bit technology. It is perhaps natural to assume that 16 bits must be better than 1 bit, but clearly it is not so simple. One point I guess may be important is that single-bit is inherently highly linear, but a 16 bit converter needs accurately trimmed components to achieve the same linearity. In a 25 year old player could it be that some of those 'accurate' components have drifted far enough to cause noticeable effects? Anyway, I will search eBay for one of the later Technics models to compare to the Marantz. There are then various well known 'improvements' to experiment with, for example replacing the output muting transistors with a relay.
I have made one, probably final, change to the phono-preamp, which is to omit the 100k preset offset adjustments for the op-amps. For some unknown reason they had to be set close to one track end to get close to zero output, but removing them entirely there was only about 1mV output. I should maybe investigate this further, one problem I can see is that there could be some interaction via the 5k6 in the 'rumble cancellation' circuit, but anyway I don't need extremely low output dc level so I have now removed the 100k presets from the circuit diagram. I have left them in the layout diagrams, but they should be considered optional, and not recommended. I have replaced the 220R presets with fixed resistors on my own board after first finding the settings for low dc output from the input amplifier section. The presets need to be removed from the board after adjustment before measuring their values, otherwise there may be errors. Parallel pairs of resistors were used to get close to the measured preset values. I doubt whether there would be any significant improvement from replacing the presets, so this also can be regarded as optional.
The phono pre-amp is completed and preliminary testing shows that it is working well, with no signs of instability. My usual distortion measuring methods using test signal nulling are not easily applicable, but distortion testing is not really essential, the distortion will be primarily second and third harmonic, and the high feedback loop gains will ensure it is at a very low level. The RIAA accuracy, as I mentioned earlier, will depend mostly on component accuracy, so testing my own version only shows how accurate my components are, so I just did a check on the relative gains at 100Hz, 1kHz and 10kHz to be sure there are no serious errors. What is easily checked is input overload level, and the peak inputs before clipping are about 800mV at 20kHz, 200mV at 1kHz and 50mV at 100Hz, all of which are very good. Assuming a cartridge with typical output 1mV/cm/sec and referring to the Shure plot of maximum observed recorded velocities the peak input voltages likely to be encountered at the three frequencies are about 55mV, 40mV and 5mV respectively, so there is a good safety margin, particularly with my own low output cartridge. I used a 20k gain setting resistor, and in practice this should ensure sufficient overload margins even for high output cartridges.
The 100k op-amp offset adjustment presets needed to be adjusted close to the track end, which is slightly worrying, but maybe normal. The input stage presets were set for zero outputs from the two input stages, and then measured as 153R and 163R, so I will replace these with fixed resistors. There is really not much more I want to do other than connect it to an amplifier and have a listen. I have added a few photos, and there is a board layout page to help anyone wanting to try this design.
I am in the process of replacing my old pre-amp using a case from an old Cambridge ATAC3, which is similar to the A1, which can often be found for sale 'faulty' on eBay, which is how I got mine. The power amplifier was rather poor, using TO92 case devices driving the power output transistors, and when these turn to smoke they can take a whole chain of other components with them. I am only using the case, the input switching and the volume control, which is the dual-concentric type, which avoids a separate balance control. To avoid the input sockets I have soldered all the input cables direct to the input board. The original transformer was a little noisy so I replaced it with a smaller and lower voltage type, and added the +/-15V regulator shown on the phono-amp page. I will also add a headphone amplifier at some stage, not necessarily the fet circuits I already included on the website which were intended as self-contained amplifiers with active volume controls.
Future plans include updating my FM tuner, this is a Denon TU-260L, again bought 'faulty' on eBay, which just needed a slight adjustment to the detector stage inductor. This has a reasonable VHF tuner (it works well for my own reception conditions anyway) but a rather ordinary filter, detector and stereo decoder. I have some good linear phase 10.7 MHz filters which I used in an old DIY design which worked well apart from a not so good VHF tuner, which used what were then the latest dual-gate mosfets but for unknown reasons, probably instability, had low gain and high noise. A transplant operation could combine the best of both designs. Then again there are some very good old tuners, if I remember correctly Technics and Yamaha were particularly good back in the 1980s, so renovating one of those could be a good option.
I have no plans for power amplifier designs, the MJR7-Mk5 is, I believe, about the best I can do with a single pair of lateral mosfets at 100mA apart from the option of adding feedforward, which I included on the old website as the MJR9 just to show what is possible rather than because it has any serious point. The earlier feedforward output stage published in Electronics World as 'Class-B in a New Class' is maybe worth returning to some day, I have added a few new comments at the start of that page to explain more about how this works.
I have changed the phono pre-amp circuit a little. The first version had some problems including probably a big switch-on thump. The circuit has been simplified, and now the preset pot is used as first stage load. Replacing with a fixed resistor, after adjusting for zero output voltage from the input stage and measuring the value, is possibly a good idea.
I have what may be the final circuit for my RIAA phono pre-amp, and have added it as Phono Pre-amp Circuit. It includes the warning that none of the circuits shown have been built and tested, they are just theoretical designs. I have most of the components, and have started working out a board layout, so all being well it will eventually be built. Beyond stability and overload margins there is probably nothing worth measuring, the RIAA accuracy will depend almost entirely on component tolerances, so if I made an inverse RIAA filter that would have the same level of errors as the pre-amp, and so testing for an overall flat response will not be helpful. This is one example where a simulation result is good enough, an actual measured result would just show how accurate or inaccurate my particular set of components were.
About 15 years ago I checked a few published RIAA phono pre-amp designs for accuracy, and found that only 2 out of 7 had anywhere near the correct equalisation component values. That was possibly an unrepresentative sample, but recently I checked a published 'inverse RIAA' network I wanted to use for testing purposes and again found an error. I also saw a very expensive pre-amp tested by Stereophile with surprising frequency response errors, so this seems to be a recurring problem. The correct equations for one common equalisation network were published in Wireless World in 1969, and with modern simulation techniques it is easy to check for errors.
I need a pre-amp for my own use, and at first I intended to just copy a simple op-amp based circuit, but then I decided to try something more unusual. The ideas I wanted to try are nothing new, they were known at least 35 years ago, but these things get forgotten, rediscovered a few times, and even renamed, so they may not be familiar to everyone. For this reason plus the continuing equalisation errors I thought it worthwhile to add a design theory article. It ended up as 8 parts, a consequence of having a long winter holiday with nothing much else to do. Eventually I will try building and testing a real circuit, but for now it is just Phono Pre-amp Design Theory.
I really want to start adding some new material soon. I am working on an article covering various aspects of RIAA phono pre-amp design. I need a new pre-amp for my own use, so this should eventually lead to a practical design. As always I want to try something slightly unusual rather than just copy one of the many published designs.
I have almost finished rewriting and transfering to the new website. The original plan was to delete the old Angelfire site, but there is still a lot of old information which may be of interest, so I may eventually just give it a new home page with links to those pages not on the new site.
Another consideration is that searching for 'audio amplifier design' on Google lists the Angelfire site at the top of the 1st page, but the new site is just about nowhere. It seems unavoidable to keep both sites for now.
I have been experimenting with Audio DiffMaker, a program which analyses pairs of audio signals and extracts the difference. In Part 2 of my capacitor distortion article I included the 'error' introduced by two different 2u2 input coupling capacitors as a sound file, and said that to me there appeared to be very little difference. Using DiffMaker I compared the two signals, and was surprised to find that the difference signal produced is quite large, so I am not entirely convinced that the result is correct. The measurements of the peak frequency spectrum suggest that the difference is not in frequency response, and the signals don't sound vastly different, so I can only guess that it is the different phase shifts which is responsible, as expected from my simulations of the effect of D/A in Part 1. As I said before, extracting the difference is of limited help, it tells us only that a difference exists, not which is better. The simulations suggested that the 'poorer' capacitor with D/A actually reduced phase errors and transient errors. Listening to the extracted capacitor voltages I think should be the most revealing test if there really are important sonic differences.
Adding resistive losses to a capacitor can reduce the resulting phase shifts compared to an ideal purely reactive component, but in input coupling applications any phase shift is in principle an unwanted effect. Anyway, the reduced error from a 10u type compared to 2u2 suggests that the capacitance value is by far the most important parameter for input coupling.
I started to investigate further, but installing DiffMaker had the unfortunate effect of preventing my spectrum analyser program OscilloMeter from starting. I had to uninstall DiffMaker and reinstall OscilloMeter. I have not looked further into this other than repeating the installation to check that it was not just coincidence.
I have updated the home page, there are now links to articles about distortion measurement and speaker design.
I am continuing to update some older pieces and then transfer them to the new website. Re-reading old material there are always bits I think need improving or clarifying. The output stage protection article has been added and also the original 'feedforward class-B output stage'. That design continues to attract some interest, and it is probably the only really clever and original idea I ever had, so although I think the later mosfet amplifiers are better I don't like to be too discouraging. I have one small improvement I never tried, so maybe someday I will try another variation using a more ordinary circuit with LTP input stage and direct coupled output.
I have added a Capacitor Distortion Part 2 page. The plan was to extract the error added by various input coupling capacitors, but with any of the test signals tried there appeared to be no signifficant difference betwen a polyester and a polypropylene both with value 2u2. A more sensitive test could subtract the two effects to reveal any difference, but trying a 10u non-polar electrolytic gave a big reduction in the signal related voltage across the capacitor, which is what we really want to reduce. My only conclusion was that increasing the capacitor value is far more beneficial than any small difference between dielectrics, unless something unusually bad such as a high-k ceramic is used.
I don't have any immediate plans for either construction projects or articles, but I had a collection of short pieces which never became articles, so I have strung a few together in a page of 'Assorted Items'. Some are not very good, or at least too much personal opinion, and one is about DSP which I really know very little about, so maybe I got some of that wrong.
On a totally different subject, I still have a physics section hidden away on the old website, with an article about relativity including application to rotating reference frames. I was reminded of this on hearing the announcement from CERN about their apparently faster than light neutrinos. The speed measurement needed accurate distance and time measurements. To measure the time between emission and detection 730km away clocks at those two points must first be accurately synchronised. There are at least two 'correct' ways to do this (and an infinite number of incorrect ways), one is global and the other local as explained at the above rotating frames link. The two methods do not match up exactly, for example locally synchronising a series of clocks round the perimeter of a rotating disc will require a discontinuity at some point. A global method could involve transporting clocks symmetrically outwards from the centre to the perimeter, which avoids the discontinuity, and will match up with clocks synchronised in a non-rotating frame, but gives local errors because the effect of the Coriolis force makes adjacent clocks experience different gravitational potentials during transport from the centre so that they run at different rates. The use of the GPS satellites for timing suggests global synchronisation was used, but then there is a mention of checking with some sort of portable timing device, which I would assume was only valid for local synchronisation, so it is not clear to me how this was done. Anyway, I am sure these sort of things were accounted for, and even if not the resulting error should be far less than the 60nsec discrepancy found. Actual faster than light travel would be far more interesting, it is allowed in current theories for virtual particles, but for real particles it would in principle allow transmission of messages backwards through time, with consequent problems for causality. If a faster than light particle is travelling forward in time in one reference frame then there exist other reference frames in which it travels backwards in time.
I have been trying alternatives to the 2SA1209 and 2SC2911, which are becoming more difficult to buy. The 2SA1381 and 2SC3503 should be good substitutes, and I have bought some of the Fairchild versions, the KSA1381E and KSC3503D. The E gain range is 100 - 200 and the D range is 60 - 120. The higher gain types are no advantage for the cascode stage so the NPN type can be D range. I found that these are in a fully insulated version of the TO-126 case, so there is no visible metal back plate, so the thick lines on my layout diagram used to indicate the 'metal back' should be understood to refer to the plain back surface, with the printing on the front.
I have substituted these types in my own prototype, and so far have found no problems, so after a few more checks these could become the recommended types.
I think there may be a problem with my email service at present, I have had a few emails saying I haven't been heard from for a while, even though I had sent messages recently, so if anyone is having problems getting in touch I apologise. I have replied to all the emails I have received, but clearly some of my replies have gone astray. Checking spam filter settings may be a good idea for anyone having problems.
It's summer holiday time and there will be a short delay before either website is updated. I planned to add some affiliate advertising to the new site to make it self-financing, that is why there were initially large unused spaces at the sides of pages, but now that is abandoned and only the full page version appears now. I also had a plan to write a few articles aimed more at beginners. There were a few problems I remember having difficulty with myself many years ago when first taking an interest in circuit design, so maybe covering that limited range of topics would be more useful than duplicating information already available on many other sites.
I wrote a section about slew rate a while ago, but I decided to replace it with a new article, Slew Rate, to make it a little clearer and include a few diagrams. This is just standard theory, but there are a few points not always covered. I mentioned the effect of increasing input stage current for a conventional degenerated differential input stage, this increases slew rate limit and also reduces distortion at lower levels. I originally guessed that doubling the current would reduce distortion only by a factor of 2, but after more thought revised this to 4, and then did a calculation for one example and got a factor of 7.78, which makes me think this is a more complex question than I first thought. This is relevant to the article on input stage distortion where I compared a few different circuits and concluded that the CFP was the best. I mentioned there that different signal levels would change the relative distortion levels of the different circuits, but I should have mentioned that the stage current would also affect results. There seems no point going through endless variations of input stages and their operating conditions to compare distortion, there are always other conflicting factors of importance such as noise and offset currents.
The limited availability of the transistors specified for the MJR7 could be a problem, so I have been looking at alternatives. The lateral mosfets are available from a number of sources still, and there are a few alternative lateral types which are almost identical to the Renesas types, made by Exicon and Semelab, the main difference is the lack of internal gate protection zeners. Having used external zeners this is unimportant, but the usual precautions should be taken to avoid static damage. Keeping the gate and source connected with metal foil or something similar until it is firmly attached to the board should be sufficient. A relatively cheap option is the ACD102PSD p-channel and ACD100NSD n-channel from a company in the UK called class-d, who sell them for £3.00 each (Inc VAT). I have not tried their mosfets myself but someone who used them says they work ok. That company sell their own amplifier boards using these mosfets, so it is possible they will continue to have them available for some time.
The small signal transistors for the MJR7 are a problem. I have plenty of 2SC2240BL and BC560C, but Farnell don't seem to have anything obvious to substitute for the 2SA1209 or 2SC2911. In USA Mouser have the KSA1381E in large quantities at a low price, and these are rated 300V which may be some advantage. (I mentioned in my 'Common-Mode Distortion' page that high voltage transistors could be expected to suffer less from base-width modulation. That was little more than a guess, and searching on Google failed to find confirmation, but I have now seen one comparison of VAS transistors (Groner, Fig.34 page 28) where high voltage rating was found to correlate fairly well with low distortion). T5 has higher power dissipation than the others and something in the original TO126 case will be more reliable, maybe with a small heatsink if using much more than a 60V supply.